Method and apparatus for use in digitally tuning a capacitor in an integrated circuit device

ABSTRACT

A method and apparatus for use in a digitally tuning a capacitor in an integrated circuit device is described. A Digitally Tuned Capacitor DTC is described which facilitates digitally controlling capacitance applied between a first and second terminal. In some embodiments, the first terminal comprises an RF+ terminal and the second terminal comprises an RF− terminal. In accordance with some embodiments, the DTCs comprise a plurality of sub-circuits ordered in significance from least significant bit (LSB) to most significant bit (MSB) sub-circuits, wherein the plurality of significant bit sub-circuits are coupled together in parallel, and wherein each sub-circuit has a first node coupled to the first RF terminal, and a second node coupled to the second RF terminal. The DTCs further include an input means for receiving a digital control word, wherein the digital control word comprises bits that are similarly ordered in significance from an LSB to an MSB.

CROSS REFERENCE TO RELATED APPLICATIONS—CLAIMS OF PRIORITY

This patent application is a continuation application of U.S. patentapplication Ser. No. 15/279,302, filed Sep. 28, 2016 entitled “Methodand Apparatus for use in Digitally Tuning a Capacitor in an IntegratedCircuit Device”, now U.S. Pat. No. 10,382,031 issued on Aug. 13, 2019,which is a continuation of patent application is a continuationapplication of U.S. patent application Ser. No. 14/638,917, filed Mar.4, 2015 entitled “Method and Apparatus for use in Digitally Tuning aCapacitor in an Integrated Circuit Device”, now U.S. Pat. No. 9,667,227issued on May 30, 2017, which is a divisional U.S. patent applicationSer. No. 12/735,954, filed Aug. 27, 2010, now U.S. Pat. No. 9,024,700issued on May 5, 2015, entitled “Method and Apparatus for Use inDigitally Tuning a Capacitor in an Integrated Circuit Device”, which isa national stage application filed pursuant to 35 U.S.C. § 371 ofinternational application number PCT/US2009/001358 filed Mar. 2, 2009(published Sep. 3, 2009 as publication number WO/2009/108391 A1), whichapplication claims the benefit of priority to commonly-assigned U.S.Provisional Application No. 61/067,634, filed Feb. 28, 2008, entitled“Method and Apparatus for Digitally Tuning a Capacitor in an IntegratedCircuit Device”. The above-identified U.S. patent applications,provisional patent application and international application numberPCT/US2009/001358 are hereby incorporated herein in their entirety byreference.

BACKGROUND 1. Field

This invention relates to integrated circuit devices, and moreparticularly to a method and apparatus for digitally tuning thecapacitance of integrated circuit components in integrated circuitdevices.

2. Related Art

Capacitors are used extensively in electronic devices for storing anelectric charge. As is well known, generally speaking, capacitorsessentially comprise two conductive plates separated by an insulator.Capacitors are used in a plurality of electronic circuits including, butnot limited to, filters, analog-to-digital converters, memory devices,various control applications, power amplifiers, tunable (also referredto as “adaptive” or “reconfigurable”) matching networks, etc.

One well-known problem to those skilled in the art of the design andmanufacture of integrated circuits is the poor tolerance valuesassociated with integrated circuit components, especially the tolerancevalues of passive circuit components. Due to process variations, deviceparameter spread, variations in critical parameters such as conductivelayer sheet resistance values, film thickness, process uniformity andmanufacturing equipment cleanliness, and other factors, integratedcircuit passive electrical components often have tolerances that areapproximately an order of magnitude worse than their analogous discreteexternal passive electrical components. Consequently, it has provendifficult and costly in the past to implement tuned networks or circuitsusing on-chip passive electrical components.

Post-fabrication trimming techniques can be used after manufacturing andtesting an integrated circuit in order to physically alter the circuitusing a variety of methods including “Zener-zapping”, laser trimming andfuse trimming Disadvantageously, the prior art post-fabricationtechniques produce only static solutions. Although the trimmed devicesmay perform adequately under nominal conditions, they may not performadequately under all of the operating conditions of the integratedcircuit. Therefore, methods for improving the tolerances of passiveelectrical devices in an integrated circuit are needed which do notrequire the use of post-fabrication trimming techniques. Further, animproved method and apparatus is needed which dynamically monitors andcorrects the performance characteristics of integrated circuits underall operating conditions. The improved method and apparatus shouldmonitor and correct the performance characteristics of tuned networksespecially as these performance characteristics are adversely affectedby poor tolerances of on-chip passive electrical devices, and by thevariable operating conditions of the device.

FIG. 1 shows a prior art attempt at solving the problem of implementingan adaptively tuned circuit using on-chip passive electrical devices. Asshown in FIG. 1, using an integrated switchable capacitor circuit 100,two terminals of an integrated tuned circuit (i.e., terminal A 101 andterminal B 103) can be selectively coupled to a bank of switchablyconnected capacitors (C₁ through C_(n)). Each of the capacitors isselectively coupled between the terminals 101, 103 by closing anassociated and respective coupling switch S_(n) For example, capacitorC₁ 102 is coupled between the terminals 101, 103 by closing anassociated switch S₁ 110. Similarly, capacitor C₂ 104 is coupled betweenthe terminals 101, 103 by closing an associated switch S₂ 112. Finally,capacitor C_(n) 108 is coupled between the terminals 101, 103 by closingan associated switch S_(n) 116. Because the individual capacitors areconnected in a parallel configuration, the total capacitance between theterminals 101, 103 is equal to the sum of the individual capacitors thatare switched into the circuit (assuming that the switches do not alsointroduce capacitance to the circuit). By electrically connecting theterminals 101, 103 to a tuned circuit that is on the same integratedcircuit as the switchable capacitor circuit 100, the capacitors can beselectively switched in and out of the tuned circuit, thereby changingthe capacitance between the terminals 101, 103 to a desired value. Thus,despite the potentially poor tolerance characteristics of the capacitorsC₁ through C_(n), the tuned circuit can be adaptively adjusted tooperate within desired parameters by simply changing the capacitancebetween terminals A 101 and B 103.

Disadvantageously, this prior art approach is undesirable when the tunedcircuit operates at relatively high frequencies. For example, when thetuned circuit operates in the GHz range of operating frequencies, thebank of switches (e.g., 110, 112, 114, and 116) introduce significantloss into the tuned circuit and thereby degrade the circuit'sperformance characteristics. The prior art solution shown in FIG. 1 alsodisadvantageously increases both the amount of space (i.e., integratedcircuit real estate) and the amount of power required to accommodate andoperate the switches. Power requirements are increased due to the D.C.current required to operate the bank of switches.

As is well known, there is an ongoing demand in semiconductor devicemanufacturing to integrate many different functions on a single chip,e.g., manufacturing analog and digital circuitry on the same integratedcircuit die. For example, recently there have been efforts to integratethe various mobile telephone handset (or cell phone) functions andcircuits in a single integrated circuit device. Only a few short yearsago, the integration of digital baseband, intermediate frequency (IF),and radio frequency (RF) circuitry on a single System-on-Chip (SoC)integrated circuit seemed improbable or nearly impossible owing to anumber of factors such as incompatible process technologies, yieldlimitations, high testing costs, poor matching of passive components,and lack of on-chip passive components having adequate analogcharacteristics. However, a number of advancements have been made incircuit design, physical implementation of hardware components, processtechnologies, manufacturing and testing techniques. These advancementsare making the integration of digital baseband, mixed-signal and RFcircuitry into a single integrated circuit device more of a reality. Onesuch advancement is described in an article entitled “Overcoming the RFChallenges of Multiband Mobile Handset Design”, by Mr. Rodd Novak,RF/Microwave Switches and Connectors, published Jul. 20, 2007,www.rfdesign.com. This article is incorporated by reference herein as ifset forth in full.

As described in the Novak paper, the complexity of cellular telephoneshas increased rapidly, moving from dual-band, to tri-band, and morerecently, quad-band. In addition, cellular phones need to be able toaccommodate a variety of signals for peripheral radios, such asBluetooth™, Wi-Fi, and GPS. This trend is expected to continue as othercapabilities are added. As described in the Novak paper, handsets arenow being developed that incorporate tri-band WCDMA and quad-band EDGEplatforms. These architectures demand at least seven radios in a singlehandset. Complexity will continue to rise due to the increasedpopularity of peripheral radios and functions that also need access tothe antenna. The increased complexity in mobile telephone handset designhas greatly complicated the RF front-end by more than tripling thenumber of high-power signal paths. By its nature, a multiband handsetmust accommodate a plurality of RF signal paths that all operate ondifferent bandwidths. Yet, all of the RF signal paths must share accessto a single antenna. As described in the Novak paper, a very efficientsolution is to route all of the competing RF signal paths to the antennausing a single single-pole, multi-throw, RF switch.

The assignee of the present application has developed and is presentlymarketing such RF switches, and exemplary RF switch designs aredescribed in applications and patents owned by the assignee of thepresent application. For example, the following applications and patentsdescribe RF switch designs that facilitate further integration of mobilehandset circuitry: U.S. Pat. No. 6,804,502, issuing Oct. 12, 2004 toBurgener, et al., U.S. Pat. No. 7,123,898, issuing Oct. 17, 2006, alsoto Burgener, et al., (both patents entitled “Switch Circuit and Methodof Switching Radio Frequency Signals”); pending U.S. application Ser.No. 11/582,206, filed Oct. 16, 2006, entitled “Switch Circuit and Methodof Switching Radio Frequency Signals”; U.S. application Ser. No.11/347,014, filed Feb. 3, 2006, and entitled “Symmetrically andAsymmetrically Stacked Transistor Grouping RF Switch”; U.S. Pat. No.7,248,120, issuing Jul. 24, 2007 to Burgener, et al.; U.S. Pat. No.7,088,971, issuing Aug. 8, 2006 to Burgener, et al.; U.S. applicationSer. No. 11/501,125, filed Aug. 7, 2006, entitled “Integrated RF FrontEnd with Stacked Transistor Switch”; and U.S. application Ser. No.11/127,520, filed May 11, 2005, and entitled “Improved Switch Circuitand Method of Switching Radio Frequency Signals”. All of the above-notedpatent applications and issued patents are incorporated by referenceherein as if set forth in full.

While these advancements in RF switch design facilitate furtherintegration of mobile handset circuitry, a significant problem ispresented as a result of mismatched impedances present at the mobilehandset antenna terminal. Due to the variable operational environment ofthe mobile handset causing the impedance at the antenna terminal to varyover a wide range, antenna impedance mismatch poses significanttechnical challenges for the mobile handset design engineer. Theproblems associated with antenna impedance mismatch are described in apaper entitled “Antenna Impedance Mismatch Measurement and Correctionfor Adaptive CDMA Transceivers”, authored by Qiao, et al., Published12-17 Jun. 2005, by the IEEE in the 2005 Microwave Symposium Digest,2005 IEEE MTT-S International, at Pages 4 et seq. (hereafter “the Qiaopaper”), and incorporated by reference herein as if set forth in full.

As described therein, mobile handsets are used in a variety ofconfigurations and positions, by users who manipulate the handset and,in particular, the antenna, in ways that are difficult to predict. Whilea nominal antenna provides an input impedance of 50 ohms, in actualusage the impedance at the antenna terminal can vary over a wide range,characterized by a voltage standing wave ratio (VSWR) of up to 10:1.(Qiao paper, see the Abstract). Consequently, it is a major designengineering challenge to maintain proper operation of the mobile handsetover a wide range of antenna impedances.

For example, for the receiver, the non-optimal source impedance degradesnoise figure, gain and dynamic range. For the power amplifier, theantenna impedance mismatch greatly impacts the efficiency, power gain,maximum output power and linearity. In the worst case, the high standingwave amplitude or possible oscillation caused by the mismatch in thecircuit may damage the power amplifier. As described in theabove-incorporated Qiao paper, in accordance with one prior artsolution, an isolator, or Voltage Standing Wave Ratio (VSWR) protectioncircuitry, is inserted between the amplifier and the antenna in order tomitigate problems associated with the antenna impedance mismatch.Unfortunately, this solution is disadvantageous because it createsattenuation, and therefore decreases antenna efficiency. Other possiblesolutions include correcting the impedance mismatch using dynamicbiasing of the power amplifier or using a tunable matching network.Adaptively correcting for environmental changes that cause antennaimpedance variation (e.g. placing a finger on top of cellphone antenna)is an important motivation for the need for tunable components inhandset RF front-ends. In addition, tunable components also allow the RFfront-end to cover more and more frequency bands, without increasing thenumber of antennas in the cellular phone. One antenna needs to covermore frequency bands in the cellular phone. This has proven difficult toachieve in prior art mobile handsets. Using tunable matching networks,the performance of the amplifier can be preserved even under severemismatch conditions. Several examples of tunable matching networks canbe found in the prior art.

For example, exemplary tunable matching networks for use in mitigatingproblems associated with antenna impedance mismatch are described in apaper entitled “An Adaptive Impedance Tuning CMOS Circuit for ISM2.4-GHz Band”, authored by Peter Sjöblom, Published in the IEEETransactions on Circuits and Systems—I: Regular Papers, Vol. 52, No. 6,pp. 1115-1124, June 2005, (hereafter “the Sjöblom paper”). As describedtherein, adaptive (or reconfigurable) matching networks are used betweenthe RF antenna and RF switch in order to continuously adapt to thechanging antenna impedance. The adaptive matching networks described inthe Sjöblom paper are implemented using a bulk CMOS process in aconfiguration using switched capacitor banks in conjunction withinductors. The capacitors and the inductors create a ladder network. Onthe antenna side, a voltage detector is followed by an analog-to-digital(A/D) converter. A controller system controls the adaptive matchingnetwork by switching the bank of capacitors through all possiblecombinations to arrive at a state yielding the best performance. FIGS.2A and 2B show two exemplary prior art tunable matching networks (200and 200′, respectively) made in accordance with the Sjöblom teachings.As shown in FIG. 2A, an exemplary tunable matching network 200 comprisesa bank of three switched capacitors 202 coupled to an inductor 204 and aload 206. The load 206 typically comprises an RF antenna. To gain enoughlatitude to match a wide range of impedances, a single inductor will notsuffice. An alternative prior art adaptive matching network 200′ isshown in FIG. 2B. The alternative network includes two inductors (204′and 204″), and three capacitor banks (208, 210, and 212), arranged asshown in FIG. 2B, and coupled to the antenna 214. The inductors (204,204′, and 204″) are typically located in “flipchip packaging” orlow-temperature co-fired ceramic (LTCC) substrates.

Disadvantageously, the tunable networks described in the Sjöblom paperdo not, and cannot be designed to provide sufficient power required bysome wireless telecommunication applications. For example, the powerhandling capabilities of the tunable networks 200, 200′ are insufficientfor mobile handsets designed for use in the well-known Global System forMobile communications (GSM). In order to be able to be used in aGSM/WCDMA handset the tunable component needs to tolerate at least +35dBm of power without generating harmonics more than −36 dBm (based onthe GSM spec). Also the IMD3 (3^(rd) order intermodulation distortion)for WCDMA needs to be sufficiently low (typ −105 dbm . . . −99 dbm).These are the same requirements that are imposed on handset antennaswitches. The Sjöblom paper is designed for low power applications (typ+20 . . . +25 dBm). It uses a single FET and a capacitor, whereas thedigitally tuned capacitor (hereafter, “DTC”) of the present teachingsuses a stack of many FETs (typ 5-6) that improve the power handlingcapabilities of the DTC. Anything built on a bulk CMOS process cannotmeet the higher power handling requirements. The UltraCmos process hasthe ability to allow use of stack transistors in the DTC therebyallowing the DTC to handle high power levels (similar to GSM/WCDMAantenna switches). Stacked transistors cannot be implemented using abulk CMOS process due to problems associated with substrate coupling.

The above-referenced Qiao paper describes a tunable matching network 300comprising silicon-on-sapphire (SOS) switches 302 coupled to shuntcapacitors 304. An exemplary prior art tunable matching network 300 madein accordance with the Qiao teachings is shown in FIG. 3. As shown inFIG. 3, this tunable matching circuit comprises six transistors 302which provide 64 (2⁶) possible capacitor states. The best state isselected to meet any particular mismatch circumstance. The tunablematching network 300 is implemented on a PCB board using discretecomponents. The transistors 302 comprise 1000 μm*0.5 μm FETs arranged inparallel and combined by wire bonding. The ON resistance for the totalswitch is approximately 0.5 ohms, and the OFF capacitance isapproximately 1.8 pF. While the switched capacitor approach taught byQiao, et al., has promising aspects, an integrated circuitimplementation using this approach would occupy significant integratedcircuit real estate. For example, the die area estimate is approximately1.2 mm² per 0.5 ohm FET, which for a six bit switched capacitor exceeds7.2 mm² without the capacitors 204. A complete tunable matching networkrequires a total of four switched capacitor banks, leading to a totalFET area of almost 30 mm². In addition to the unwieldy die area requiredby the Qiao teachings, it is also difficult to accurately control theoverall capacitance due to the tolerance differences in the discretecapacitors. The circuit also disadvantageously has inferior powerhandling capabilities, linearity and Q-factor values for someapplications. In addition, in this prior art solution, degradation inperformance is caused by parasitic inductance of discrete capacitors. Itis advantageous to use integrated capacitors (as opposed to discretecapacitors) because the parasitic inductance and Quality-factor (Q) ofan integrated solution is higher using an integrated circuit on asapphire substrate than what is typically achievable using discrete SMDcapacitors.

As described in both the above-referenced Qiao and Sjöblom papers, athigher frequencies using integrated circuit technology, much work hasbeen done using Micro-Electromechanical Systems (MEMS) switches insteadof CMOS switches and capacitors. MEMS switches, varactors and thin-filmBarium Strontium Titanate (BST) tunable capacitors have been used in thedesign of tunable or switched matching networks. Disadvantageously,these approaches have disadvantages of cost, tuning range (also referredto as “tuning ratio”) (which generally corresponds with maximumavailable capacitance/minimum available capacitance), integration andlinearity. For various reasons, these solutions fail to meet the powerhandling, tuning ratio, and linearity requirements imposed by manywireless telecommunication specifications. Even after years of researchand development, several MEMS and BST manufacturing enterprises thatwere founded to pursue the tunable component opportunities have fallenshort of the requirements and specifications set forth in variouscellular telephone specifications. Consequently, mass produced tunablecapacitors or inductors for GSM power levels (i.e., +35 dBm) and WCDMAlinearity (IMD3 −105 dBm) simply do not exist. BST capacitors exhibitsignificant problems when operated at high temperatures where theirQ-factor is significantly degraded.

For example, varactor diodes and bulk CMOS switched capacitors do notmeet the power and linearity requirements of these cellularspecifications. MEMS switched capacitor banks exist, but they do notseem to meet power and linearity requirements, they require separatehigh-voltage driver chip and hermetic packaging, and reliability is aproblem in mobile handset applications. BST voltage tunable capacitorsare based on ferroelectric materials. These prior art solutions havedifficulty meeting power and linearity requirements. They alsodisadvantageously require an external high voltage (HV) integratedcircuit in order to produce high bias voltages (e.g., 20-40V) andgenerally cannot be integrated with other control electronics. The BSTvoltage tunable capacitors also suffer from degraded performances due tohysteresis and temperature stability.

Therefore, a need exists for a method and apparatus for digitally tuninga capacitor in an integrated circuit device. A need exists for a methodand apparatus that can overcome the disadvantages associated with theprior art solutions and that facilitates the integration of tunablecapacitor networks on a single integrated circuit. The need exists foran apparatus that facilitates the full integration of a tunable matchingnetwork for use with other mobile handset circuits and functions. Inaddition, the need exists for an apparatus and method that candynamically calibrate an integrated tuned capacitor network such as atunable antenna matching network. The present teachings provide such amethod and apparatus.

The details of the embodiments of the present disclosure are set forthin the accompanying drawings and the description below. Once the detailsof the disclosure are known, numerous additional innovations and changeswill become obvious to those skilled in the art.

SUMMARY

A method and apparatus for use in a digitally tuning a capacitor in anintegrated circuit device is described. A Digitally Tuned Capacitor DTCis described which facilitates digitally controlling capacitance appliedbetween a first and second terminal. In some embodiments, the firstterminal comprises an RF+ terminal and the second terminal comprises anRF− terminal. In accordance with some embodiments, the DTCs comprise aplurality of sub-circuits ordered in significance from least significantbit (LSB) to most significant bit (MSB) sub-circuits, wherein theplurality of significant bit sub-circuits are coupled together inparallel, and wherein each sub-circuit has a first node coupled to thefirst RF terminal, and a second node coupled to the second RF terminal.The DTCs further include an input means for receiving a digital controlword, wherein the digital control word comprises bits that are similarlyordered in significance from an LSB to an MSB. Each significant bit ofthe digital control word is coupled to corresponding and associatedsignificant bit sub-circuits of the DTC, and thereby controls switchingoperation of the associated sub-circuit. DTCs are implemented using unitcells, wherein the LSB sub-circuit comprises a single unit cell. Nextsignificant bit sub-circuits comprise x instantiations of the number ofunit cells used to implement its associated and corresponding previoussignificant bit sub-circuit, wherein the value x is dependent upon aweighting coding used to weight the significant bit sub-circuits of theDTC. DTCs may be weighted in accordance with a binary code, thermometercode, a combination of the two, or any other convenient and useful code.In many embodiments, the unit cell comprises a plurality of stacked FETsin series with a capacitor. The unit cell may also include a pluralityof gate resistors R_(G) coupled to the gates of the stacked FETs, and aplurality of R_(DS) resistors coupled across the drain and source of thestacked FETs. The stacked FETs improve the power handling capabilitiesof the DTC, allowing it meet or exceed high power handling requirementsimposed by current and future communication standards.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified schematic of a prior art attempt at solving theproblem of implementing an adaptively tuned circuit using on-chippassive electrical devices.

FIGS. 2A and 2B are schematics of exemplary prior art tunable matchingnetworks comprising banks of shunt capacitors coupled to respectiveinductors and a load.

FIG. 3 is a schematic of another exemplary prior art tunable matchingcircuit comprising a stack of six transistors which provide 64 (2⁶)possible capacitor states.

FIG. 4A is a simplified schematic representation of one embodiment of adigitally tuned capacitor (DTC) in accordance with the presentteachings.

FIG. 4B is a simplified schematic representation of the DTC of FIG. 4A.

FIG. 4C is a simplified schematic representation of another embodimentof a digitally tuned capacitor (DTC) in accordance with the presentteachings.

FIG. 4D is a simplified schematic representation of the DTC of FIG. 4C.

FIG. 5A is a simplified schematic representation of another embodimentof a digitally tuned capacitor (DTC).

FIG. 5B is an equivalent circuit showing the ON resistances and OFFcapacitances associated with the switching FETs of the DTC of FIG. 5A.

FIG. 5C is a simplified schematic representation of another embodimentof a digitally tuned capacitor (DTC).

FIG. 5D is an equivalent circuit showing the ON resistances and OFFcapacitances associated with the switching FETs of the DTC of FIG. 5C.

FIG. 6A shows design details of another embodiment of a DTC made inaccordance with the present teachings; wherein the DTC is designed inaccordance with a unit cell design block technique, and wherein the DTCincludes a plurality of stacking FETs coupled in series with associatedand corresponding MIM capacitors.

FIG. 6B shows design details of another embodiment of a DTC made inaccordance with the present teachings; wherein the DTC comprises a moregeneralized version of the DTC of FIG. 6A and is designed in accordancewith a unit cell design block technique, and wherein the DTC includes aplurality of stacking FETs coupled in series with associated andcorresponding MIM capacitors.

FIG. 7A is a schematic of a generalized unit cell design block (an LSBsub-circuit) that is used to implement a DTC in accordance with thepresent teachings.

FIG. 7B is a schematic of an ON state RF equivalent circuit of the unitcell design block of FIG. 7A.

FIG. 7C is a schematic of a simplified equivalent circuit of the ONstate RF equivalent circuit of FIG. 7B.

FIG. 7D is a plot showing the Q vs. freq curve for the unit cell designblock of FIGS. 7A-7C.

FIG. 7E is a schematic of an OFF state RF equivalent circuit of the unitcell design block of FIG. 7A.

FIG. 7F is a schematic of a simplified equivalent circuit of the OFFstate RF equivalent circuit of FIG. 7E.

FIG. 7G is a plot showing the QOFF vs. freq for the OFF state RFequivalent circuit of FIG. 7E.

FIG. 7H shows a simplified equivalent circuit of a fully implemented andcomplete DTC using the design principles and concepts described withreference to FIGS. 7A-7G.

FIG. 7I shows a simplified equivalent circuit of the fully implementedand complete DTC of FIG. 7H.

FIG. 7J shows a simplified schematic of a FET stack showing how aneffective FET stack height is achieved using the present teachings,wherein the effective stack height exceeds the actual stack height ofthe present DTC.

FIG. 8A is a schematic of an exemplary 1 GHz DTC made in accordance withthe design characteristics set forth in Table 1.

FIG. 8B shows a model simulation of the 1 GHz DTC of FIG. 8A.

FIG. 8C is a plot of the total capacitance of the DTC of FIG. 8A versusthe DTC capacitance control word setting.

FIG. 8D is a plot of the total Q-factor value versus the DTC capacitancecontrol word setting of the DTC of FIG. 8A for a given applied signalfrequency.

FIG. 8E shows an exemplary integrated circuit layout of a 1×bit LSB unitcell of the DTC of FIG. 8A made in accordance with the presentteachings.

FIG. 8F shows an exemplary integrated circuit layout of the 1 GHz DTC ofFIG. 8A.

FIG. 9A is a schematic of an exemplary 2 GHz DTC made in accordance withthe design characteristics set forth in Table 1.

FIG. 9B shows a model simulation of the 2 GHz DTC of FIG. 9A.

FIG. 9C is a plot of the total capacitance of the DTC of FIG. 9A versusthe DTC capacitance control word setting.

FIG. 9D is a plot of the total Q-factor value versus the DTC capacitancecontrol word setting of the DTC of FIG. 9A for a given applied signalfrequency.

FIG. 9E shows an exemplary integrated circuit layout of a 1×bit LSB unitcell of the DTC of FIG. 9A made in accordance with the presentteachings.

FIG. 9F shows an exemplary integrated circuit layout of the 2 GHz DTC ofFIG. 9A.

FIGS. 10A and 10B show a comparison of the capacitance tuning curves ofthe present DTCs with those of thin-film Barium Strontium Titanate (BST)tunable capacitors.

FIG. 11 shows a graph of the tuning range of a DTC versus the frequencyof the applied signal for a selected minimum Q-factor value (Q_(min)).

FIG. 12 shows a graph of the tuning range and die area requirementsversus minimum Q-factor values (Q_(min)) for a selected DTC at a givenapplied signal frequency of 900 MHz.

FIG. 13 shows plots showing how FET die area requirements (i.e., the diearea requirement of the FETs of the DTC) associated with different FETstack heights increases as the maximum DTC capacitance (Cmax) increases.

FIG. 14A is a graph showing plots of the tuning ranges and die arearequirements versus minimum Q-factor values (Q_(min)) for a selectedunmodified DTC.

FIG. 14B is a graph showing plots of the tuning ranges and die arearequirements versus minimum Q-factor values (Q_(min)) for a modifiedDTC, wherein the modified DTC comprises the unmodified DTC modified toinclude a fixed capacitor in parallel thereto.

FIG. 15A is a simplified schematic of an unmodified DTC made inaccordance with the present teachings, and wherein FIG. 15A alsoincludes DTC parameter values.

FIG. 15B is a simplified schematic of a modified DTC made in accordancewith the present teachings, wherein the modified DTC is implemented bycoupling a fixed capacitor in parallel with the unmodified DTC of FIG.15A.

FIG. 15C shows a simplified schematic of a four terminal ACC MOSFET madein accordance with “HaRP” design techniques, wherein the ACC MOSFET isused to implement FETs comprising the FET stack in some embodiments ofthe DTC.

Like reference numbers and designations in the various drawings indicatelike elements.

MODES OF CARRYING OUT THE INVENTION

Throughout this description, the preferred embodiment and examples shownshould be considered as exemplars, rather than as limitations on thepresent invention.

FIG. 4A shows a simplified schematic representation of a one embodimentof a digitally tuned capacitor (hereafter, “DTC”) 400 for use in anintegrated circuit device in accordance with the present teachings. Asshown in FIG. 4A, in one exemplary embodiment, the DTC 400 comprises aplurality of capacitors (e.g., capacitors 402, 404, 406 and 408) havingfirst terminals coupled in series to respective MOSFET devices (i.e.,capacitor 402 is coupled to the source of FET 402′, capacitor 404 iscoupled to the source of FET 404′, capacitor 406 is coupled to thesource of FET 406′, and capacitor 408 is coupled to the source of FET408′). In the embodiment shown in FIG. 4A, second terminals of thecapacitors 402, 404, 406 and 408 are coupled to a ground node or groundterminal 410. However, in a more general implementation of a DTC made inaccordance with the present teachings, the second terminals of thecapacitors 402, 404, 406 and 408 may be coupled together and coupled toan ungrounded terminal or port. Such ungrounded terminal or port may, insome embodiments, be coupled to a load, an RF port or terminal (anegative or positive RF port), or to any other convenient port orterminal.

As shown in FIG. 4A, the drains of the FETs are coupled together, andcoupled to a load terminal 412. Thus, the load terminal 412 and groundterminal 410 are analogous to the terminals A 101 and B 103 of FIG. 1,respectively. As described in more detail below, in some embodiments,the load terminal 412 may comprise a mobile handset antenna. As shown inthe more generalized DTC 400″ of FIGS. 4C and 4D, the load terminal 412comprises an RF+ terminal 412′, and the “ground” terminal 410 (which is,in the more generalized case described below with reference to FIGS. 4Cand 4D, not necessarily coupled to ground at all) comprises an RF−terminal 410′. These more generalized embodiments of the DTC aredescribed in more detail below. Those skilled in the electronic devicedesign arts shall recognize that the plurality of capacitors (402-408)may alternatively be coupled to the drains of the FETs (402′-408′), andthe sources may be coupled to the load terminal 412, depending onwhether the FETs comprise N-type of P-type MOSFETs. In one embodiment inaccordance with the present teachings, the plurality of capacitorscomprise metal-insulator-metal MIM capacitors. As is well known, MIMcapacitors are widely used in monolithic integrated circuits inDC-decoupling, matching, and biasing circuits. In integrated circuitdevices, the various MIM capacitors advantageously exhibit very goodmatching characteristics (i.e., they have excellent tolerancecharacteristics).

Although the DTCs of the present teachings are described throughout thepresent application as being implemented using MIM capacitors (e.g., thecapacitors 402-408 of FIG. 4A), it will be appreciated by those skilledin the electronic design arts that the MIM capacitors may, in otherembodiments, comprise different capacitor types. More specifically,these capacitors may comprise any useful RF capacitor having a highQ-factor value. In some embodiments, the capacitors may comprise MIM(Metal-Insulator-Metal), MM (Metal-Metal), Interdigitated Capacitors(IDC) and their variants. The “MIM” capacitors may also, in otherembodiments, comprise FETs biased in an OFF state.

In accordance with the present teachings, the capacitance values of theMIM capacitors (i.e., the capacitors 402-408) are weighted in aconvenient and desirable manner. For example, in one embodiment, the MIMcapacitors of the DTC are given a binary weighting. More specifically,in accordance with this embodiment, the least-significant capacitor C₁402 is designed to have a desired least significant (or lowest)capacitance of C_(LSB). The next significant capacitor C₂ 404 isdesigned to have a capacitance of twice C_(LSB), or 2*C_(LSB). Thebinary weighting is assigned in like fashion with each next significantcapacitor having a capacitance that is a power of two greater than theprevious significant capacitor. Finally, the most significant capacitorC_(n) 408 is designed to have a capacitance of 2^(n-1)*C_(LSB).

Those skilled in the IC manufacturing arts will appreciate that severalalternative means may be used to implement the capacitance of a selectedcapacitor. For example, in one embodiment, the selected capacitor (e.g.,C₂ 404) can be formed by placing two previous significant capacitors (inthis example, C₁ 402) in parallel. Similarly, the next significantcapacitor (e.g., C₃ 406) can be formed by placing four of the leastsignificant capacitors (e.g., C₁ 402) in parallel. Alternatively, thecapacitors may be designed to different physical dimensions to have thedesired capacitance values. In addition, although the MIM capacitors ofthe embodiment shown in FIG. 4A are described as having a binaryweighting, those skilled in the electronic design arts shall recognizethat any convenient capacitance-weighting scheme can be assigned to theMIM capacitors. For example, in an alternative embodiment wherein alogarithmic scaling is desired, each capacitor can be designed to have acapacitance value that is ten times greater than its previoussignificant capacitor. More specifically, and referring again to FIG.4A, capacitor C₂ 404 can be designed to have a capacitance that is10*C_(LSB), wherein C₁ 402 is designed to have a capacitance of C_(LSB).In this embodiment, C_(n) is assigned a capacitance of 10^(n-1)*C_(LSB).As described below in more detail, in accordance with one embodiment ofthe present teachings, the MIM capacitors are weighted using a“thermometer coding” scheme.

As described in more detail below, in one embodiment of the present DTC,the MIM capacitors (e.g., the capacitors 402-408) are designed as partof a “unit cell” design block. As described in more detail below, theunit cell comprises a fundamental design building block that can bereplicated (or instantiated) within an integrated circuit device toachieve a desired function. In accordance with the unit cellimplementation, the least significant capacitor (i.e., capacitor C₁ 402)is part of a unit cell design block. For example, the unit cell designblock may comprise the least significant bit (LSB) sub-circuit whichcomprises the least significant FET 402′ coupled in series with theleast significant shunt capacitor C₁ 402 (shown in FIG. 4A as a “unitcell design block” 414). In accordance with the unit cellimplementation, the capacitance of a selected MIM capacitor (e.g., thesecond least significant shunt capacitor C₂ 404) comprises two unit cellblocks 414 electrically coupled in parallel. That is, the nextsignificant bit sub-circuit comprises two instantiations of the unitcell design block (which comprises the LSB sub-circuit as describedabove). The capacitance of the next significant bit capacitor (i.e., C₃406) comprises four unit cell blocks 414 electrically coupled inparallel, and so on. The MSB significant bit sub-circuit comprises 8instantiations of the LSB sub-circuit, coupled in parallel. Thetolerances and matching of the MIM capacitors (402-408) are greatlyimproved using the unit cell design approach because they are based onidentical unit cell building blocks. This implementation is described inmuch more detail below.

In accordance with one embodiment of the present DTC, both thecapacitance values of the MIM capacitors (e.g., MIM capacitor C₁ 402)and the size of their respective FETs (e.g., FET 402′) are weightedsimilarly. For example, and referring again to FIG. 4A, the leastsignificant FET 402′ can be designed to comprise the smallest (i.e., FEToccupying the least integrated circuit die area) FET of the plurality ofFETs used in the DTC 400. FET sizes are dimensioned such that the Qspecification is met (Ron of the FET vs the Cmim capacitance) and alsoso that a desired tuning ratio is achieved. The capacitance of the FETwhen it is turned OFF is represented by “Coff”. So when the FET is OFF,the total capacitance of the bit is C_(mim) in series with C_(OFF). Theselection of the FET size and thus C_(OFF) of each FET determines theCmin, or minimum capacitance for the entire DTC. Also, in a stack ofFETs there is voltage division between the FETs. The MIM capacitor valuecan also be adjusted such the required stack height of FETs can bereduced, based on voltage division between C_(OFF) of the FETs andC_(mim). Owing to its smallest size, the least significant FET 402′therefore has the highest ON Resistance (R_(ON), which is defined hereinas the resistance of the FET when it is turned ON) and the lowest OFFCapacitance (C_(OFF), which is defined herein as the capacitance of theFET when it is turned OFF) as compared to all of the other FETs (e.g.,404′, 406′ and 408′) of the DTC 400. For example, in one embodiment, ifthe least significant bit FET 402′ has an ON resistance of R_(ON), andan OFF capacitance of C_(OFF), the next significant bit FET 404′ can bebinary weighted (similar to the binary weighting of the MIM capacitors)to be twice the size of its previous significant bit FET (i.e., 402′),and therefore have an ON resistance of R_(ON)/2, and an OFF capacitanceof C_(OFF)*2. Similarly, the next significant bit FET 406′ is binaryweighted to be four times the size of the least significant FET (i.e.,FET 402′), and therefore have an ON resistance of R_(ON)/4, and an OFFcapacitance of C_(OFF)*4.

The binary weighting of the FETs are assigned in like fashion (similarto the binary weighting of the MIM capacitors) with each nextsignificant bit FET having an ON resistance that is half that of theprevious significant bit FET, and an OFF capacitance that is twice thatof the previous significant bit FET. Finally, the most significant bitFET (e.g., the FET 408′ of the DTC 400) FET_(n) is designed to have asize that is 2^(n-1)*FET_(LSB) (wherein n is the number of FETs used inthe DTC). In this embodiment, the most significant FET has a size thatis 2^(n-1)*the size of the least significant bit FET. The mostsignificant bit FET therefore has an OFF capacitance that is2^(n-1)*COFF_(LSB) (wherein COFF_(LSB) comprises the C_(OFF) of theleast significant bit FET), and an ON resistance that isRON_(LSB)/2^(n-1) (wherein RON_(LSB) comprises the ON resistance of theleast significant bit FET). As described above, similarly to theweighting of the MIM capacitors, other weighting schemes can be appliedto the FETs. For example, a thermometer weighting scheme can be used.However, in the general case, whatever weighting scheme is used, itshould be applied equally to both the MIM capacitors and theirrespective and associated FETs. For example, if a binary weightingscheme is used, it should be applied to each corresponding significantbit FET and MIM capacitor, on a one-to-one basis. Whatever weight isassigned to a selected capacitor (e.g., the MIM capacitor C₃ 406) shouldalso be assigned to its corresponding and associated FET (i.e., the FET406′). This configuration is described in more detail below. This aspectof the present DTC teachings is important because it maintains constantQ values for each of the bits. Constant Q factors are maintained for theFETs because the relationship between Ron and Cmim stays the same due tothe scaling aspect. This also causes the Q-factor value of the entireDTC to remain the same as the unit cell (assuming all FETs are turnedON).

Because the plurality of MIM capacitors are coupled together in parallelas shown in FIG. 4A, their respective capacitance values combine bysimply adding the capacitance values of all of the individual MIMcapacitors. The capacitance of the DTC 400 (as measured between the load412 and ground 410) is therefore equal to the sum of the capacitance ofall of the MIM capacitors C_(n).

Referring again to FIG. 4A, the capacitance between the load 412 andground terminal 410 (i.e., the total capacitance of the DTC 400) iscontrolled by a digital control word CAP_(word) 426 that is applied to acontrol logic block 416. In some embodiments, the control wordCAP_(word) 426 is applied directly to the DTC FETs without use of anintervening control logic block 416. The control word that is applied tothe DTC FETs may be generated using a feedback circuit that identifiesand tracks operation of a mobile telephone handset (for example, it maybe continuously generated by monitoring impedance matching of the mobilehandset with a handset antenna and adjusting the control wordaccordingly). Those skilled in the electronics design arts shallrecognize that there are many ways to generate the digital control wordin order to control the capacitance of the DTC 400, and such mechanismsare contemplated by and fall within the scope of the present teachings.

Referring again to FIG. 4A, the control word is applied to individuallycontrol the switching operation of each of the FETs (i.e., 402′-408′) ofthe DTC 400. The control bits are ordered from least significant bit(LSB) to most significant bit (MSB), and are assigned to control theshunting FETs associated and corresponding to the least significant MIMcapacitor to the most significant MIM capacitor. The least significantbit (e.g., B₀) of the control word is applied on signal line 418 tocontrol the operation of the least significant bit FET 402′. The nextsignificant bit (e.g., B₁) of the control word is applied on signal line420 to control the operation of the next significant bit FET 404′. Thenext significant bit (e.g., B₂) of the control word is applied on signalline 422 to control the operation of the next significant bit FET 406′.Finally, the most significant bit (e.g., B₃) of the control word isapplied on signal line 424 to control the operation of the mostsignificant bit FET 408′. In the example shown in FIG. 4A, a four bitcontrol word controls the operation of the four FETs, therebycontrolling which (and how many) of the MIM capacitors are appliedbetween the load terminal 412 and ground 410. In the DTC 400 shown inFIG. 4A, the DTC can have one of sixteen (i.e., 2⁴) possible discretecapacitance values. FIG. 4B is a simplified schematic representation ofthe DTC 400 shown in FIG. 4B.

FIGS. 4C and 4D show simplified schematic representations of generalizedembodiments of digitally tuned capacitors (DTCs) 400″, and 400″′,respectively, made in accordance with the present teachings. Thegeneralized embodiment of the DTC 400″ of FIG. 4C functions similarly tothe DTC 400 described above with reference to FIG. 4A. However, as shownin FIGS. 4C and 4D, the generalized DTC 400″ (and generalized DTC 400″′of FIG. 4D) digitally tunes or varies the capacitance between a first RFterminal (specifically, an RF+ terminal 412′) and a second RF terminal(specifically, an RF− terminal 410′). The sign designations shown in theDTCs of FIGS. 4C and 4D, and associated with the first and second RFterminals (i.e., the “+” and “−” sign designations), merely indicate atop terminal (i.e., “RF+” 412′) and a bottom terminal (i.e., “RF−” 410′)of the generalized DTCs 400″ and 400″′. The RF+ terminal 412′ isanalogous to the terminal A 101 of the prior art switchable capacitorcircuit 100 of FIG. 1. The RF− terminal 410′ is analogous to theterminal B 103 of the prior art switchable capacitor circuit 100 ofFIG. 1. The RF+ 412′ and RF− 410′ terminals of the DTC 400″ may becoupled to any convenient port, terminal, load, or other circuit device,as required to meet design parameters and system requirements.

For example, in some embodiments the DTC 400″ is coupled to othercircuits in a “Shunt” configuration. When coupled in such a “shunt”configuration, the RF+ terminal 412′ may be coupled to a load or RF portand the RF− terminal 410′ may be coupled to ground (i.e., connectedsimilarly to connection of the DTC 400 described above with referenceFIG. 4A). In another embodiment of a shunt configuration, the RF+terminal 412′ may be coupled to ground and the RF− terminal 410′ may becoupled to a load or RF port. In still further embodiments, the DTC 400″may be coupled to other circuits in a “Series” configuration. Whencoupled in a “series” configuration the RF+ terminal 412′ may be coupledto an input port, such as, for example, an RF input port, and the RF−terminal 410′ may be coupled to an output port, such as, for example, anRF output port. In another embodiment of a series configuration, the RF+terminal 412′ may be coupled to an output port, such as, for example, anRF output port, and the RF− terminal 410′ may be coupled to an inputport, such as, for example, an RF input port.

The DTC 400″ of FIG. 4C also shows the plurality of MIM capacitors ascoupled in series at the top of a stack of FET switches. Thisconfiguration is described in more detail below. FIG. 4D is a simplifiedschematic representation of the DTC 400″ of FIG. 4C. The DTC 400″ alsoshows an implementation of a “5-bit” DTC, wherein the digital controlword applied to control the tuning of the DTC comprises 5 bits, and theDTC 400″ is therefore implemented using 5 significant bit sub-circuits.As described above, in accordance with one embodiment of the unit celldesign technique of the present teachings, each significant bitsub-circuit is implemented by coupling an appropriate number of unitcells together in parallel. For example, the LSB significant bitsub-circuit comprises the unit cell. The next significant bitsub-circuit comprises two instantiations of the unit cell, coupled inparallel. The next significant bit sub-circuit comprises fourinstantiations of the unit cell, also coupled in parallel. Finally, asshown in FIG. 4C, the MSB significant bit sub-circuit comprises 16 unitcells (or 16 LSB sub-circuits) coupled in parallel.

FIG. 5A shows a simplified schematic representation of anotherembodiment of a digitally tuned capacitor (DTC) 500 for use in anintegrated circuit device in accordance with the present teachings. Asshown in FIG. 5A, in one exemplary embodiment, the DTC 500 comprises aplurality of capacitors coupled in series to a plurality of switchingshunt FETs 504. Note that the plurality of MIM capacitors 502 of the DTC500 of FIG. 5A are coupled between the plurality of shunt FETs 504 andan RF antenna terminal 506 (i.e., the MIM capacitors 502 are coupled “ontop” of the shunt FETs 504 as contrasted with being coupled below theshunt FETs). Also, as described above with reference to the generalizedDTC 400″ and as described below in more detail below with reference tothe generalized DTC 500″ of FIG. 5C, a generalized implementation of theDTC 500 facilitates digitally tuning of the capacitance between a firstterminal and a second terminal of the DTC. That is, although the DTC 500is shown in FIGS. 5A and 5B as having an RF antenna terminal 506 (shownin FIGS. 5A and 5B as coupled to a first terminal of the MIM capacitors502) and as having a ground terminal 510 (shown in FIGS. 5A and 5B asbeing coupled to the bottom (or drains) of the shunt FETs 504, ageneralized implementation of a DTC is not so limited. As described inmore detail below with regard to the more generalized DTC 500″ of FIG.5C (and DTC 500′″ of FIG. 5D), the RF antenna terminal 506 of the DTC500 of FIG. 5A (and the RF antenna terminal 506 of the DTC 500′ of FIG.5B) may comprise an RF+ terminal 506′. The “ground” terminal 510 (whichis, in the more generalized case described below with reference to FIGS.5C and 5D, not necessarily coupled to ground at all) may comprise an RF−terminal 510′. These embodiments are described in more detail below withreference to the more generalized DTC 500″ of FIG. 5C and the DTC 500′″of FIG. 5D.

The 5-bit DTC 500 (control word bits b_(o) through b₄ are used tocontrol the total capacitance of the DTC 500) functions similarly to the4 bit version described above with reference to FIGS. 4A and 4B. Anequivalent circuit 500′ is shown in FIG. 5B showing the ON resistancesand OFF capacitances associated with the shunt FETs 504 of FIG. 5A. Therelative capacitances of the MIM capacitors 502 are also shown in FIG.5B. As shown in FIG. 5B, and similar to the DTC 400 of FIG. 4A, the DTC500′ of FIG. 5B uses a binary weighting scheme. Specifically, the leastsignificant bit (LSB) FET 504′ has an ON resistance of R_(ON) and an OFFcapacitance of C_(OFF). Its associated and corresponding MIM capacitor502′ has a capacitance of C_(MIM). The next significant bit FET 504″ hasan ON resistance of R_(ON)/2 and an OFF capacitance of 2C_(OFF). Itsassociated and corresponding MIM capacitor 502″ has a capacitance of2C_(MIM). The remainder of the DTC 500′ is similarly binary weighted,with the most significant bit FET 504″″ having an ON resistance ofR_(ON)/16 and an OFF capacitance of 16C_(OFF). Its associated andcorresponding MIM capacitor 502″″ has a capacitance of 16C_(MIM).Although not shown in FIG. 5B, each of the MIM caps in reality have aninherent loss term that is associated with it. The MIM Q value isapproximately 100.200. The inherent loss would be represented by aresistor shown in series with the MIM.

As noted in the description of FIG. 5B, the Q-factor (or Quality factor)of each significant bit sub-circuit of the DTC 500′ (i.e., the ONresistance R_(ON) and C_(MIM) values for each of the MIMcapacitor/shunting FET sub-circuits [e.g., the LSB FET 504′ coupled inseries with its corresponding and associated MIM capacitor 502′] shownin FIG. 5B) are identical. In addition, the total Q-factor of the DTC500′ is identical to the Q-factor of each sub-section of the DTC 500′when all of the shunting FETs (i.e., when all of the FETs 504′, 504″,504′″ . . . 504″″) are turned on. As is well known, the Q-factor, or“Factor of Merit”, of a device is a measure of “quality” of that device.It is often used to indicate the efficiency of a device or circuit (forexample, it can be used to compare the frequency at which a systemoscillates to the rate at which it dissipates its energy). As is wellknown, many of the present wireless telecommunication specificationsimpose strict Q-factor requirements on RF front end circuitry. Forexample, the RF front end circuitry must exhibit low loss and have aQ-factor typically in the 50-100 range.

In one embodiment, as described above with reference to the DTC 400 ofFIG. 4A, the Q (or Quality-factors) of each sub-circuit section (i.e.,the values of R_(ON) and C_(MIM)) are identical because the DTC 500′ isimplemented using the above described unit cell design technique. Asdescribed above, in accordance with this design technique, an LSBsignificant bit sub-circuit 503′ (i.e., defined herein as the LSB FET504′ coupled in series with the LSB MIM capacitor 502′) comprises a unitcell design block. All next significant bit sub-circuits (e.g., the nextsignificant bit sub-circuit 503″ comprising the FET 504″ and itsassociated and corresponding MIM capacitor 502″) are implemented byinstantiating (or replicating) the LSB sub-circuit 503′ (which comprisethe unit cell design block for the described DTC 500′) as many times asrequired to achieve binary weighting. For example, the LSB sub-circuit503′ (LSB FET 504′ coupled in series with the LSB MIM capacitor 502′) isinstantiated twice (i.e., it is replicated), and the two instantiationsare coupled in parallel, to implement the next significant bitsub-circuit 503″ (comprising the FET 504″ and its associated andcorresponding MIM capacitor 502″). The LSB sub-circuit is instantiatedfour times (and coupled in parallel) to implement the next significantbit sub-circuit comprising the FET 504″′ and its associated andcorresponding MIM capacitor 502″′, and so on. Finally, as shown in FIG.5B, the MSB significant bit sub-circuit 503″″ comprises 16instantiations of the LSB sub-circuit (and coupled in parallel).

In accordance with one embodiment of the present DTC method andapparatus, the DTC is designed in accordance with the followingidealized design equations (Equations 1-4):

$\begin{matrix}{{C_{\min} = {\left( {2^{bits} - 1} \right)\frac{C_{MIM} \cdot C_{OFF}}{C_{MIM} + C_{OFF}}}};} & {{Equation}\mspace{14mu} 1} \\{{C_{\max} = {\left( {2^{bits} - 1} \right)C_{MIM}}};} & {{Equation}\mspace{14mu} 2} \\{{{{Tuning}\mspace{14mu}{ratio}} = {{C_{\max}/C_{\min}} = {1 + \frac{C_{MIM}}{C_{OFF}}}}};} & {{Equation}\mspace{14mu} 3} \\{{Q_{\min} = {\frac{1}{\omega\; C_{MIM}R_{ON}} = \frac{1}{{\omega \cdot \left( {{C_{\max}/C_{\min}} - 1} \right) \cdot R_{ON}}C_{OFF}}}};} & {{Equation}\mspace{14mu} 4}\end{matrix}$

wherein Cmin comprises the minimum capacitance that can be produced bythe DTC 500′, Cmax comprises the maximum capacitance that can beproduced by the DTC 500′, “bits” represents the number of bits in thecontrol word, Tuning ratio (also referred to herein as “Tuning range”)comprises the range of capacitances over which the DTC can be tuned, andwherein Qmin comprises the minimum allowable Q factor of the DTC 500′.As those skilled in the electronics design arts shall recognize, inpractice, the “non-ideal” Q-value of the MIM capacitors would need to beaccounted for in Equation 4 above. However, Equation 4 comprises an“idealized” equation, so this non-ideal Q factor is not accounted fortherein.

As noted briefly hereinabove, FIGS. 5C and 5D show generalizedimplementations of the DTC 500 (of FIG. 5A) and 500′ (of FIG. 5B),respectively. The DTCs shown in FIGS. 5C and 5D function similarly totheir respective DTC counterpart implementations, with the followingimportant caveat. The DTC 500″ of FIG. 5C (and the DTC 500″ of FIG. 5D)includes both an RF+ terminal 506′ and an RF− terminal 510′. Asdescribed above with reference to the DTCs of FIGS. 4C and 4D, and asshown in the DTCs of FIGS. 5C and 5D, the DTC 500″ (and DTC 500″ of FIG.5D) digitally tunes or varies the capacitance between a first RFterminal (specifically, the RF+ terminal 506′) and a second RF terminal(specifically, the RF− terminal 510′). The sign designations shown inthe DTCs of FIGS. 5C and 5D, and associated with the first and second RFterminals (i.e., the “+” and “−” sign designations), merely indicate atop terminal (i.e., “RF+” 506′) and a bottom terminal (i.e., “RF−” 510′)of the generalized DTCs 500″ and b 500′″. The RF+ terminal 506′ isanalogous to the terminal A 101 of the prior art switchable capacitorcircuit 100 of FIG. 1, and it is also analogous to the RF+ terminal 412′of the DTC 400″ and 400′″. The RF− terminal 510′ is analogous to theterminal B 103 of the prior art switchable capacitor circuit 100 of FIG.1, and it is also analogous to the RF− terminal 410′ of the DTC 400″ and400′″. The RF+ 506′ and RF− 510′ terminals of the DTC 500″ (and the DTC500″ of FIG. 5D) may be coupled to any convenient port, terminal, load,or other circuit device, as required to meet design parameters andsystem requirements. In all other respects, the DTCs 500″ and 500″′ areimplemented and operate similarly to their counterpart “grounded” DTCimplementations of FIGS. 5A and 5B, and therefore no further descriptionof these DTC implementations is set forth herein.

In one embodiment, the DTCs of the present teachings are implementedusing UltraCMOS™ process technology. UltraCMOS™ comprises mixed-signalprocess technology that is a variation of silicon-on-insulator (SOI)technology on a sapphire substrate offering the performance of GalliumArsenide (“GaAs”) with the economy and integration of conventional CMOS.This technology delivers significant performance advantages overcompeting processes such as GaAs, SiGe BiCMOS and bulk silicon CMOS inapplications where RF performance, low power and integration areparamount. This process technology is described in detail in severalU.S. patents owned by the assignee of the present invention, including(but not limited to) U.S. Pat. No. 5,416,043, issuing on May 16, 1995;U.S. Pat. No. 5,492,857, issuing on Feb. 20, 1996; U.S. Pat. No.5,572,040, issuing on Nov. 5, 1996; U.S. Pat. No. 5,596,205, issuing onJan. 21, 1997; U.S. Pat. No. 5,600,169, issuing on Feb. 4, 1997; U.S.Pat. No. 5,663,570, issuing on Sep. 2, 1997; U.S. Pat. No. 5,861,336,issuing on Jan. 19, 1999; U.S. Pat. No. 5,863,823, issuing on Jan. 26,1999; U.S. Pat. No. 5,883,396, issuing on Mar. 16, 1999; U.S. Pat. No.5,895,957, issuing on Apr. 20, 1999; U.S. Pat. No. 5,930,638, issuing onJul. 27, 1999; U.S. Pat. No. 5,973,363, issuing on Oct. 26, 1999; U.S.Pat. No. 5,973,382, issuing on Oct. 26, 1999; U.S. Pat. No. 6,057,555,issuing on May 2, 2000; U.S. Pat. No. 6,090,648, issuing on Jul. 18,2000; U.S. Pat. No. 6,667,506, issuing on Dec. 23, 2003; U.S. Pat. No.7,088,971, issuing on Aug. 8, 2006; U.S. Pat. No. 7,123,898, issuing onOct. 17, 2006; and U.S. Pat. No. 7,248,120, issuing on Jul. 24, 2007.The above-cited present assignee owned patents are incorporated byreference herein as if set forth in full.

Implementing the DTCs of the present disclosure using the UltraCMOS™process technology yields the following benefits and advantages ascompared with the prior art tunable capacitor solutions: Binary-weightedswitch FETs and MIM capacitors; Linear tuning curve; GSM/WCDMA compliantpower handling (+35 dBm) and linearity (IMD3<−105 dBm) (this particularaspect is described in more detail below with reference to the figuresthat follow; also, it should be noted that this benefit is achievabledue to the stacking FETs configuration, such stacking of FETs is notpossible in bulk CMOS and is difficult in SOI implementations; however,it can be achieved using the present DTC teachings implemented inUltraCMOS, SOI and GaAs implementations); Integrated MIM capacitors,very good matching between the different MIM capacitors; No hysteresis(vs. BST solutions); No capacitance modulation with high power RF signal(vs. BST solutions); Standard control logic and VDD voltages (vs.BST/MEMS); Fast switching speed (approximately 1-3 μS); Highreliability, manufacturability (vs. BST and MEMS prior art approaches);Flip-chip packaging option for low parasitic inductance; and Scaledback-end technology reduces the die area by 40%.

Although the DTC of the present application is described as beingimplemented in the above-cited UltraCMOS process technology, thoseskilled in the electronics arts shall appreciated that the DTC of thepresent teachings can also be implemented in any convenient integratedcircuit process technology including, but not limited to,Silicon-on-Insulator (SOI) CMOS and GaAs process technology.

FIG. 6A shows another embodiment of a DTC 600 made in accordance withthe present teachings. The DTC 600 of FIG. 6A teaches the use ofstacking FETs which is necessary to meet high power requirements imposedby system standards. Nominally, in one exemplary embodiment, one FET canwithstand Max_Vds=+2.54V RF voltage across its source and drain. Notethat the specified Vds voltage across the FET refers to the RMS valueand not the peak value of the voltage. In order to handle GSM powerlevels, in one embodiment the DTC would use a stack height of seven. Thevoltage handling in this example, then is equal to 7*2.54V=17.8V. The RFpower handling in 50 ohm can be calculated based upon this value. When aMIM capacitor is placed on top of and in series with the FET stack,additional capacitive voltage division between Cmim and Coff of each FEToccurs. If Cmim had the identical value as Coff, the stack height can bereduced by one FET (i.e., a stack-of-6 FETs plus one MIM, instead ofstack-of-7). If the Cmim is smaller or larger than Coff, the effectivepower handling for the DTC can be calculated such that max_Vds (i.e.,the maximum voltage that any FET in the FET stack can withstand) is notexceeded for each FET. The MIM capacitors can withstand much highervoltages than the FETs.

The embodiments of the present DTCs shown in FIGS. 6A and 6B also teachdesign techniques including scaling of FETs, MIM capacitors, R_(DS) andR_(G) resistors to achieve the desired DTC functions. While the DTCsdescribed above with reference to FIGS. 4A, 4C, and 5A-5D comprisesimplified implementations, the DTCs of FIGS. 6A and 6B show moredetailed and practical DTC implementations. As shown in FIG. 6A, forexample, the DTC 600 comprises a plurality of stacked switching FETscoupled in series with associated and corresponding MIM capacitors. Forexample, in one embodiment, a least significant bit (LSB) sub-circuit602 comprises a plurality of shunting FETs (in the examples shown inFIG. 6A the plurality comprises six shunting FETs) arranged in a stackedconfiguration, and coupled in series with a MIM capacitor 604. Thestacked FETs (i.e., the FETs 606, 608, 610, 612, 614 and 616) arecoupled together in series, and, in turn, the FET stack is coupled inseries with the MIM capacitor 604. In one embodiment, the stacked FETsare implemented in accordance with a U.S. patent and pending patentapplications owned by the assignee of the present patent application.More specifically, in accordance with this embodiment, the stacked FETs(e.g., the FETs 606-616) are implemented in accordance with U.S. Pat.No. 7,248,120, entitled “Stacked Transistor Method and Apparatus,”issued to Burgener, et al., on Jul. 24, 2007; or in accordance withpending U.S. patent application Ser. No. 11/347,014, entitled“Symmetrically and Asymmetrically Stacked Transistor Grouping RFSwitch”, filed Feb. 3, 2006 in the name of Kelly, et al., or inaccordance with pending U.S. patent application Ser. No. 11/501,125,entitled “Integrated RF Front End with Stacked Transistor Switch”, filedAug. 7, 2006 in the name of Burgener, et al. The above-cited U.S. patent(U.S. Pat. No. 7,248,120) and applications (application Ser. No.11/347,014 and application Ser. No. 11/501,125) are incorporated byreference herein as if set forth in full.

As described in the above-incorporated patent and pending applications,the FET stacking configuration increases the power handling capabilitiesof the DTC 600. By increasing the number of stacked transistors in thestacked transistor groupings (i.e., by increasing the stacked FET“height”), the DTC 600 is able to withstand applied RF signals havingincreased power levels. The stacked FET configuration allows the DTC 600to meet the stringent power handling requirements imposed by the GSM andWCDMA wireless telecommunication specifications. For example, the GSMand WCDMA specifications require power handling of approximately +35dBm. Stacking the shunt FETs as shown in the least significant bit (LSB)sub-circuit 602 allows the DTC 600 to meet the high power handlingrequirements of the GSM and WCDMA specifications. The MIM capacitor 604also drops some of the voltage across it which allows a reduction in therequired FET stack height (i.e., it allows less stacked FETs to be usedin order to meet the desired power handling requirements of the DTC600).

In other embodiments, the least significant bit (LSB) sub-circuit 602further includes a plurality of gate resistors (R_(G)) coupled to thegates of the stacked FETs and the least significant bit (b₀) of thecontrol word. In these embodiments, the LSB sub-circuit 602 alsoincludes a plurality of drain-to-source resistors (R_(DS)) configured asshown, wherein each R_(DS) is coupled across the drain and source of itsassociated and corresponding shunting FET, and wherein the R_(DS)resistors are coupled in series between the MIM capacitor 604 and aground node 618. As described below in more detail with reference to themore generalized DTC 600′ of FIG. 6B, the ground node 618 may beimplemented as an RF− terminal (terminal 618′ of FIG. 6B). The gateresistors (R_(G)) and drain-to-source resistors (R_(DS)) are requiredfor biasing their associated and corresponding shunting FET devices.More specifically, the R_(G) resistors are required as a consequence ofthe stacked FET configuration. Without stacking (i.e., stack“height”=1), the R_(G) resistor could be eliminated. The R_(DS) resistoris used with the “HARP” implementation described below in more detail.However, these resistors reduce the OFF-state Q-factor of the DTC 600.Larger gate resistors (R_(G)) and drain-to-source resistors (R_(DS)) canbe used in order to improve the OFF-state Q-factor values.Unfortunately, increasing the size of these resistors also increases theintegrated circuit die area occupied by the DTC 600. The switching timeassociated with the shunting FETs is also increased thereby.

Similar to the DTC 400 and 500′ described above with reference to FIGS.4A and 5B, respectively, in one embodiment, the DTC 600 is implementedusing a unit cell design technique. Each significant bit sub-circuit ofthe DTC 600 is binary weighted similar to the binary weighting describedabove with reference to the DTCs 400 through 500′. As described above,in accordance with this design technique, the LSB sub-circuit 602comprises a unit cell design block. As described above, in someembodiments the unit cell design block (i.e., the LSB sub-circuit 602)comprises at least the LSB stacked FETs 606-616, inclusive, coupled inseries with the LSB MIM capacitor 604. In other embodiments, the unitcell design block also comprises the gate (R_(G)) resistors anddrain-to-source (R_(DS)) resistors coupled as shown in the LSBsub-circuit 602 of FIG. 6.

As described above, in the embodiment of the DTC 600 shown in FIG. 6A,the LSB sub-circuit 602 comprises a unit cell design block. All nextsignificant bit sub-circuits (e.g., the next significant bit sub-circuitcontrolled by the next significant bit b₁ of the control word) areimplemented by instantiating (or replicating) the LSB sub-circuit 602 asmany times as required to achieve the binary weighting. For example, theLSB sub-circuit 602 is instantiated twice (i.e., it is replicated), andcoupled in parallel, to implement the next significant bit sub-circuit.The LSB sub-circuit is instantiated four times (and coupled in parallel)to implement the next significant bit sub-circuit (which is controlledby the next significant bit of the control word), and so on. Finally, asshown in FIG. 6A, the MSB most significant bit sub-circuit (which iscontrolled by the most significant bit (MSB) [b_((b-1))] of the controlword, wherein “b” comprises the number of bits of the control word) isimplemented by instantiating (or replicating) the LSB sub-circuit 6022^(b-1) times.

In the embodiments of the DTC 600 wherein the unit cell design block(i.e., the LSB sub-circuit 602) comprises only the stacked FETs (i.e.,the FETs 606-616, inclusive) coupled in series with the MIM capacitor604), while the LSB sub-circuit is instantiated as described above inimplementing the next significant bit sub-circuits, the R_(DS) and R_(G)resistors are not so instantiated (or duplicated). Rather, in theseembodiments (as shown in the DTCs 600 and 600′ of FIGS. 6A and 6B,respectively), the R_(DS) and R_(G) resistors are scaled in half foreach successive significant bit sub-circuit. For example, As shown inFIG. 6A, although the MIM capacitors (e.g., the MIM capacitors 604, 620,622) are weighted similarly to the weighting of the analogous MIMcapacitors of the DTC 500, 500′ of FIGS. 5A and 5B, respectively, thegate resistors (R_(G)) and drain-to-source resistors (R_(DS)) havedecreasing values (for increasing significant bit sub-circuits) similarto the ON resistances (R_(ON)) described above with reference to the DTC500′ of FIG. 5B. For example, the resistance of the gate resistors(R_(G)) of the sub-circuit that is controlled by control bit b₁ is ½that of the resistance of the gate resistors of the LSB sub-circuit 602.Similarly, the resistance of the drain-to-source resistors (R_(DS)) ofthe sub-circuit that is controlled by control bit b₁ is ½ that of theresistance of the drain-to-source resistors (R_(DS)) of the LSBsub-circuit 602. The R_(G) and R_(DS) resistors for the next significantbit sub-circuits are weighted similarly. These embodiments of the DTC600 (of FIG. 6A) and the DTC 600′ (of FIG. 6B) significantly reduce theamount of integrated circuit die area required to implement the DTCs,and improves performance characteristics of the DTCs.

As noted briefly above, in another embodiment the DTC can be implementedin accordance with a thermometer weighting scheme. In accordance withthis thermometer weighting embodiment, instead of binary weighting eachof the successive significant bit sub-circuits (as implemented in theDTC 600 of FIG. 6A and the DTC 600′ of FIG. 6B), a “thermometer coding”scheme is used, wherein the entire DTC comprises 2^(n)-1 (31 for a 5 bitcapacitance control word) identical unit cell design blocks (i.e., theLSB sub-circuit 602). In the thermometer coded embodiments of the DTC,the DTC has 2^(n) possible capacitance tuning states using 2^(n)-1identical unit cell design blocks. For example, if the digital controlword comprises 5 bits, the thermometer coded embodiments of the DTC areimplemented using 31 identical unit cell design blocks and have 32possible capacitance tuning states.

The thermometer weighting advantageously results in a DTC havingidentical capacitance steps (i.e., the capacitance differentialresulting between two adjacent states of the control word, such asbetween “00000” and “00001”) and guaranteed monotonicity. In contrast,when a binary weighting scheme is used, different sized sub-circuits areswitched ON and OFF depending on which state the DTC is in. For example,when switching between a capacitance control word of 01111 and 10000,the largest (MSB) sub-circuit is turned ON, and all other significantbit sub-circuits are turned OFF. If the capacitance tolerance isrelatively poor, this can result in varying capacitance steps ascompared to, for example, switching from 10000 to 10001. Onedisadvantage with using thermometer weighting is related to the physicalsizes of the R_(DS) and R_(G) resistors. The 1×bit (LSB) unit cellcomprises the largest sized R_(DS) and R_(G) resistors. Consequently,these resistors occupy a significant portion of integrated circuit diearea. In contrast, the MSB bit sub-circuit occupies 1/16th of the areaoccupied by the 1×bit (LSB) unit cell. Consequently, implementing theDTC using thermometer weighting wastes much of the precious integratedcircuit die area due to the space occupied by the R_(DS) and R_(G)resistors. In other embodiments, it is also possible to use acombination of binary weighting and thermometer coding, or any otherconvenient weighting scheme. The DTC of the present teachingscontemplate use of any convenient weighting scheme, and theseimplementations fall within the scope and spirit of the presentteachings.

Note that the MIM capacitors (i.e., the MIM capacitors 604, 620 and 622)are positioned on top of the stack of shunting FETs as shown in FIGS. 6Aand 6B. From an RF perspective, the control lines (e.g., the controllines 640, 642 and 644) behave as if they are coupled to ground. Due tothis aspect of the DTC 600 of FIG. 6A, it is better to position the MIMcapacitors at the top, rather than at the bottom, of the FET stack inthis implementation. When the MIM capacitors are positioned at thebottom of FET stack, the R_(G) resistors are effectively placed inparallel with the MIM capacitors when the corresponding and associatedFET is in the ON state. This configuration (placing the MIM capacitorsat the bottom of their respective FET stacks) thereby reduces theirassociated Q-factor values. That being said, the present DTC teachingscontemplate use of either configuration (i.e., MIM capacitors placed ontop or bottom of the FET stack), and any such designs fall within thescope and spirit of the present DTC teachings.

Note that the MIM capacitors (i.e., the MIM capacitors 604, 620 and 622)are based on identical unit cells and therefore have excellent tolerancecharacteristics and matching between the different capacitors. Inaddition, the larger sized stacked FETs (i.e., those having more“fingers”) have smaller ON resistances (R_(ON)) and larger OFFcapacitance values (C_(OFF)) as compared to the smaller sized shuntingFETs. The stacked FETs (i.e., the FETs 606-616) of the LSB sub-circuit602 comprise the smallest sized FETs of the DTC 600. The LSB sub-circuit602 also includes the smallest sized MIM capacitor, largest gateresistors (R_(G)) and largest drain-to-source resistors (R_(DS)). Theswitching time of the stacked FETs ((the gate resistance R_(G))*(thegate capacitance C_(GATE) of the FET)) are constant across all of theFETs in the DTC 600. In addition, the ON state Q-factor of the unit cellstack (i.e., the unit cell design block 602 of FIG. 6A) is dominated bythe ON resistance (R_(ON)) of the stacked FETs (606-616) and the MIMcapacitor 604 capacitance C_(MIM).

FIG. 6B shows a more generalized version 600′ of the DTC 600 describedabove with reference to FIG. 6A. As shown in FIG. 6B, the generalizedDTC 600′ includes a first RF terminal (an RF+ terminal 680) and a secondRF terminal (an RF− terminal 618′). The RF+ terminal 680 is coupled to afirst terminal of each of the MIM capacitors of each significant bitcell (for example, it is coupled to a first terminal of the MIMcapacitor 604′ of a least significant bit (LSB) sub-circuit 602′ asshown in FIG. 6B. It is also coupled to first terminals of each of theother MIM capacitors as shown in FIG. 6B. The RF− terminal 618′ iscoupled as shown to the bottom FETs of the FET stacks of eachsignificant bit sub-circuit of the DTC (i.e., it is coupled to thedrains of the bottom FET of each FET stack). The RF− terminal 618′therefore supplants the ground terminal 618 of the DTC 600 describedabove with reference to FIG. 6A. The DTC 600′ is also generalized in thesense that it allows for any desired number of stacked FETs to be usedto implement the LSB sub-circuit 602′ (whereas the DTC 600 of FIG. 6Auses six stacked FETs). In all other respects, the DTC 600′ of FIG. 6Bis implemented and functions similarly to the DTC 600 described abovewith reference to FIG. 6A, and it is therefore not described in moredetail herein.

Similarly to the operation of the DTCs described above with reference tothe DTCs of FIGS. 4A-5D, a digital control word is applied to the DTCs600 and 600′ to selectively control the switching operation of each ofthe significant bit sub-circuits of the DTCs 600 and 600′. The controlword bits are ordered from a least significant bit (LSB) (i.e., b₀) to amost significant bit (MSB) (i.e., b_((b-1)), wherein b comprises thenumber of control word bits). As shown in FIGS. 6A and 6B, eachsignificant bit of the control word is coupled to an associated andcorresponding significant bit sub-circuit. For example, as shown in FIG.6A, the LSB b₀ of the control word is coupled via the gate resistorsR_(G) to the gates of the LSB sub-circuit 602 FET stack (i.e., it iscoupled to control the gates of the stacked FETs 606-616). The nextsignificant bit (i.e., b₁) of the control word is similarly coupled viathe gate resistors R_(G)/2 to the gates of the next significant bitsub-circuit FET stack, thereby controlling the switching operation ofthe next significant bit sub-circuit, and so on. Finally, the MSBb_((b-1)) of the control word is coupled via the gate resistorsR_(G)/2^(b-1) to the gates of the MSB significant bit sub-circuit FETstack, thereby controlling the switching operation of the MSBsub-circuit.

In one embodiment of the DTCs 600 and 600′, the FET stacks are turned ON(e.g., the stacked FETs 606-616 of the LSB sub-circuit 602 are switchedto an ON state) by applying a positive voltage at their associated andcorresponding control bits (e.g., LSB control bit b₀ 640). For example,in one exemplary embodiment, the control bits apply a positive voltageof +2.75 volts to turn ON their associated and corresponding FET stacks.Although many prior art examples use 0V (i.e., ground) to turn OFF FETdevices, in order to achieve improved linearity, the presentimplementation turns OFF the FET stacks by applying a negative voltageon their associated and corresponding control bits. For example, in oneexemplary embodiment, the control bits apply a −3.4V signal to turn OFFtheir associated and corresponding FET stacks. The more negative thecontrol voltage is, the better the linearity characteristics of the FETsin the FET stacks. However, the applied control bit voltage should notbe allowed to become too negative as it might then exceed the maximumvoltage limits of the FETs used in implementing the FET stacks. In someembodiments, the negative voltages are generated by a Negative VoltageGenerator which may be integrated on the same integrated circuit die asare the DTCs.

In addition to the Negative Voltage Generator noted above, theintegrated circuit die within which the DTCs are implemented may alsoinclude Serial Interfaces and ESD protection circuits. The DTCs may, insome embodiments, be coupled to any and all of these devices, thusallowing for the integration of additional functions on the same die asthe DTC. In addition, a single integrated circuit die may containmultiple DTCs, and the DTCs may be coupled to any and all of themultiple DTCs to achieve desired circuit and system requirements. Insome embodiments, the multiple DTCs are completely separate andunconnected to each other. Alternatively, the multiple DTCs may beconfigured in a series shunt configuration. Further, in otherembodiments, the DTCs may all be configured in a shunt configuration.

FIG. 7A is a circuit schematic showing design details of a generalizedunit cell design block 700 that is analogous to the unit cell (i.e., theleast significant bit (LSB) sub-circuit 602 described above withreference to FIG. 6A, and more accurately, the LSB sub-circuit 602′described above with reference to the DTC 600′ of FIG. 6B). As describedabove in more detail, the unit cell design block is used to implementmany embodiments of the DTC in accordance with the present teachings. Asshown in FIG. 7A, one embodiment of the unit cell 700 comprises a stackof n shunting FETs 702, wherein the stack 702 is coupled in series witha MIM capacitor 704. The individual shunting FETs of the FET stack(i.e., the FETs 706, 708, 710, 712, 714 and 716) are coupled together inseries, and the entire FET stack is coupled in series to a firstterminal of the MIM capacitor 704. A second terminal of the MIMcapacitor 704 is shown coupled to a resistor, shown in FIG. 7A asR_(MIM) 705. In this embodiment, the resistor R_(MIM) 705 comprises anEquivalent Series Resistance (ESR) of the MIM capacitor 704. The MIMcapacitor 704 is depicted in FIG. 7A as being coupled to a first RFterminal (i.e., an RF+ terminal 780) via its associated R_(MIM)reisistor 705.

The unit cell 700 also includes n drain-to-source resistors (R_(DS))configured as shown, wherein each R_(DS) is coupled across the drain andsource of an associated and corresponding shunting FET, and wherein theR_(DS) resistors are coupled in series between the first terminal of theMIM capacitor 704 and a second RF terminal (i.e., an RF− terminal 718).The unit cell 700 also includes n gate resistors R_(G) coupled to thegates of their associated and corresponding switching FETs. The n gateresistors R_(G) are coupled together at a node 720, wherein the node 720is controlled by a control bit (e.g., the LSB control bit b₀ 722 of thecontrol word). In most embodiments, the operation of the unit cell iscontrolled by the LSB control bit of the control word as shown in FIG.7A. As described above with reference to the DTC 600 and DTC 600′ ofFIGS. 6A and 6B, respectively, the unit cell is used to implement theremaining significant bit sub-circuits of the DTC. Using the unit cellapproach, the tolerances and values of the various components are verywell matched, if not identical.

As described above with reference to the DTC 600 of FIG. 6A, the gateresistors (R_(G)) and drain-to-source resistors (R_(DS)) are requiredfor biasing their associated and corresponding shunting FET devices.However, these resistors reduce the OFF-state Q-factor value of the DTC.Larger gate resistors (R_(G)) and drain-to-source resistors (R_(DS)) canbe used in order to improve the OFF-state Q-factor values.Unfortunately, increasing the size of these resistors also increases theintegrated circuit die area required to implement the DTC.

FIG. 7B shows a schematic of an ON state RF equivalent circuit 700′ ofthe unit cell 700 described above with reference to FIG. 7A. The ONstate RF equivalent circuit 700′ comprises the state of the unit cell700 wherein all of the shunting FETs (i.e., the shunt FETs 706-716,inclusive) are turned ON (i.e., wherein the LSB control bit b₀ 722 is ina state that causes all of the shunt FETs to turn ON). FIG. 7C shows aschematic of a simplified equivalent circuit 700″ of the ON state RFequivalent circuit 700′ of FIG. 7B. As shown in FIGS. 7B and 7C, whenthe unit cell 700 is in an ON state, the equivalent resistance betweenthe RF+ 780 and RF− 718 terminals comprises n*R_(ON) (i.e., the ONresistance R_(ON) of one of the FETs in the stack), added to the R_(MIM)705 resistor value (i.e., the Equivalent Series Resistance of the MIMcapacitor 704). The FET on-resistance (R_(ON)) and R_(MIM) 705 determinethe Q-factor value of the unit cell stack when the unit cell 700 is inan ON state (i.e., the “Q_(ON)” of the unit cell). As shown in the graph730 of FIG. 7D, the Q_(ON) value is proportional to 1/freq (wherein“freq” comprises the frequency of the signal applied to the unit cell).The Q_(ON) of the unit cell (shown along the y axis of the graph 730)decreases as the frequency (freq) of the applied signal (shown along thex axis of the graph 730) increases. In some embodiments, the ONresistance R_(ON) is selected to meet the minimum Q-factor specification(Qmin 732) at the highest operating frequency (F_(max), also referred toherein as “f_(MAX)”, 734) of the applied signal. Equation 5a, set forthbelow, shows the mathematical relationship of ON-state Q_(ON), f(frequency), C_(MIM), R_(MIM), n, R_(G) and R_(ON), when driving RF+terminal while RF− terminal is coupled to ground (preferred).

$\begin{matrix}{Q_{ON}^{+} = \frac{1}{2\pi\;{f \cdot {C_{MIM}\left( {R_{MIM} + \frac{1}{\frac{1}{n \cdot R_{ON}} + \frac{n}{R_{G}}}} \right)}}}} & {{Equation}\mspace{14mu} 5a}\end{matrix}$

Equation 5b, set forth below, shows the mathematical relationship ofON-state Q_(ON), f, C_(MIM), R_(MIM), n, R_(G) and R_(ON), when drivingthe RF− terminal and while the RF+ terminal is coupled to ground. Inthis case the Q value is degraded as the R_(G) is now effectively inparallel with C_(MIM). Hence it is preferred that the RF− terminal iscoupled to ground, instead of the RF+ terminal.

$\begin{matrix}{Q_{ON}^{-} = \frac{1}{2\pi\;{f \cdot {C_{MIM}\left( {\frac{n}{{R_{G}\left( {2\pi\;{f \cdot C_{MIM}}} \right)}^{2}} + R_{MIM} + {n \cdot R_{ON}}} \right)}}}} & {{Equation}\mspace{14mu} 5b}\end{matrix}$

As will be appreciated by those skilled in the electronic design arts,the equations set forth above (i.e., Equations 5a and 5b) allows the DTCto be designed to meet a given Q-factor that is required by a systemspecification or standard (e.g., as required by a wirelesstelecommunications standard such as WCDMA). For a given Q-factor (i.e.,for a given Q_(ON) value) and a given maximum operating frequency (i.e.,for a given f_(MAX)), the DTC designer may determine the C_(MIM),R_(MIM), n, R_(G) and R_(ON) values for the unit cell in accordance withEquations 5a and 5b.

FIG. 7E shows a schematic of an OFF state RF equivalent circuit 700′ ofthe unit cell 700 described above with reference to FIG. 7A. The OFFstate RF equivalent circuit 700′ comprises the state of the unit cell700 wherein all of the shunting FETs (i.e., the stacked shunting FETs706-716, inclusive) are turned OFF (i.e., wherein the LSB control bit b₀722 is in a state that causes all of the shunting FETs to turn OFF).FIG. 7F shows a schematic of a simplified equivalent circuit 700″″ ofthe OFF state RF equivalent circuit 700″′ of FIG. 7E. As shown in FIGS.7E and 7F, when the unit cell 700 is in an OFF state, the equivalentresistance between the first RF terminal (i.e., the RF+ terminal 780)and the second RF terminal (i.e., the RF− terminal 718) is determined bymany factors, including the following: R_(MIM) (the Equivalent SeriesResistance (ESR) of the MIM capacitor 704); n*R_(DS) (i.e., the totalresistance value of all of the drain-to-source resistors R_(DS) coupledin series); n*R_(COFF), wherein n comprises the number of FETs in thestack and wherein R_(COFF) comprises an Equivalent Series Resistance(ESR) of the FET OFF capacitance C_(OFF); and the value of (3/n)*R_(G),wherein R_(G) comprises the resistance value of a gate resistor.

The resistance values of the drain-to-source and gate resistors, R_(DS)and R_(G), respectively, aid in determining the Q-factor value of theunit cell stack when the unit cell 700 operates in an OFF state (i.e.,the “Q_(OFF)” of the unit cell 700). As shown in the graph of FIG. 7G,for example, for applied signal frequencies ranging from DC to a minimumFrequency (F_(min)), the Q_(OFF) value is approximately linearlyproportional to the frequency (shown as “freq” in the graph of FIG. 7G)of the signal applied to the unit cell 700. As shown in the graph ofFIG. 7G, the Q_(OFF) value increases approximately linearly as thefrequency applied to the unit cell 700 increases from DC to F_(min). TheQ factor then “flattens” or levels out as the applied signal frequencyincreases beyond F_(min). In some embodiments, the drain-to-sourceR_(DS) and gate R_(G) resistance values are selected to meet the minimumQ-factor specification (Qmin 732′) at the lowest operating frequency(F_(min), also referred to herein as “f_(MIN)”, 734′) of the appliedsignal. Equation 6a, set forth below, shows the mathematicalrelationship of OFF-state Q_(OFF), f (frequency), C_(MIM), R_(MIM), n,R_(G), R_(DS), C_(OFF) and R_(COFF), when driving the RF+ terminal andwhile the RF− terminal is coupled to ground.

$\begin{matrix}{Q_{OFF}^{+} = \frac{\frac{1}{C_{MIM}} + \frac{n}{C_{OFF}}}{2\pi\;{f \cdot \left( {\frac{n^{3}}{3{R_{G}\left( {2\pi\;{f \cdot C_{OFF}}} \right)}^{2}} + \frac{n}{{R_{DS}\left( {2\pi\;{f \cdot C_{OFF}}} \right)}^{2}} + R_{MIM} + {n \cdot R_{COFF}}} \right)}}} & {{Equation}\mspace{14mu} 6a}\end{matrix}$

Equation 6b, set forth below, shows the mathematical relationship ofOFF-state Q_(OFF), f (frequency), C_(MIM), R_(MIM), n, R_(G), R_(DS),C_(OFF) and R_(COFF), when driving the RF− terminal and while couplingthe RF+ terminal to ground.

$\begin{matrix}{Q_{OFF}^{-} = \frac{\frac{1}{C_{MIM}} + \frac{n}{C_{OFF}}}{2\pi\;{f \cdot \left( {\frac{n}{3{R_{G}\left( {2\pi\;{f \cdot C_{MIM}}} \right)}^{2}} + \frac{n}{{R_{DS}\left( {2\pi\;{f \cdot C_{OFF}}} \right)}^{2}} + R_{MIM} + {n \cdot R_{COFF}}} \right)}}} & {{Equation}\mspace{14mu} 6b}\end{matrix}$Exemplary component values for the equations set forth above comprisethe following:

R_(ON)=2.14 Ω

R_(MIM)=12.6 Ω

C_(MIM)=100e-15 F

R_(DS)=277e3 Ω

R_(G)=1106e3 Ω

C_(OFF)=280e-15 F

R_(COFF)=7 Ω

n=6

As will be appreciated by those skilled in the electronic design arts,the equation set forth above (i.e., Equation 6) allows the DTC to bedesigned to meet a given Q-factor that is required by a specification(e.g., as required by a wireless telecommunications standard such asWCDMA). For a given Q-factor (i.e., for a given Q_(OFF) value) and agiven minimum operating frequency (i.e., for a given f_(MIN)), the DTCdesigner may select C_(MIM), R_(MIM), C_(OFF), R_(COFF), n, R_(DS), andR_(G) values for the unit cell. Typically R_(DS) is set equal toR_(G)/n, wherein n comprises the stack height (i.e., the number of FETsin the stack). However, those skilled in the electronic design artsshall recognize that other values can be selected for the gate anddrain-to-source resistors without departing from the scope or spirit ofthe present disclosure.

As described above with reference to FIGS. 4A, 5A-5D, 6A and 6B, and7A-7G, DTCs implemented in accordance with the present disclosed unitcell design technique provide significant advantages as compared withthe prior art tunable capacitor solutions. Because the DTC is fabricatedusing the LSB sub-circuit as the unit cell, and because the unit cellcomprises a fundamental building block, and because all othersignificant bit sub-circuits comprise replicated versions of thisfundamental building block, the tolerances and Q-factors of the varioussub-circuits (and their components) are very well matched, and, in somecases, are identical. This aspect is in stark contrast with the priorart solutions such as the tunable matching circuits described above withreference to FIGS. 1-3, wherein the tolerances and Q-factors of theswitched capacitor circuits were not well matched, and clearly notidentical. The unit cell technique also advantageously facilitates ascalable design that can be replicated to achieve almost any tuningratio. The stacked FET configuration comprising a stack of n FETs allowsthe DTC to meet a desired power handling specification. The DTC designercan adjust n accordingly to meet the power handling requirements. If theDTC needs to handle less power, the number of FETs in the stack can bedecreased (thereby saving precious integrated circuit real estate). Incontrast, if the power handling capabilities of the DTC need to beincreased, n may also be increased accordingly. As described above withreference to FIGS. 7A-7G, the various components and electricalcharacteristics of the unit cell can be selected by the DTC designer toaccommodate almost any desired Q-factor for a given operating frequencyrange.

FIG. 7H shows a simplified equivalent circuit of a fully implemented andcomplete DTC 790 using the design principles and concepts describedabove with reference to FIGS. 7A-7G. The equivalent circuit of thecompleted DTC 790 is produced using the generalized unit cell 700 designblock described above with reference to FIG. 7A, and by coupling theequivalent circuits described above with reference to FIGS. 7B-7C and7E-7F in a manner that reflects the OFF and ON states of eachsignificant bit sub-circuit. Specifically, the fully implemented andcomplete equivalent circuit 790 of FIG. 7H is created by using theequivalent circuits described above with reference to FIGS. 7C and 7F.As indicated by switch arrows shown in FIG. 7H, either of theabove-described ON or OFF equivalent circuits are coupled between theRF+ terminal 780 and RF− terminal 718. This switching is controlled bythe digital control word applied to the DTC 790. Depending on whether aselected significant bit is turned ON or OFF, either one of thesecircuits is placed between RF+ and RF− terminals, as shown by the arrow“switch” arrow symbols.

FIG. 7I shows a simplified equivalent circuit 792 of the fullyimplemented and complete DTC 790 described above with reference to FIG.7H. The simplified equivalent circuit 792 teaches how to derive acomplete and accurate equivalent circuit useful in modeling the completeDTC 790 across all possible tunable states of the DTC. For example, fora 5 bit DTC 790, the DTC 790 may have 32 distinct tunable states. Thatis, such a DTC can produce 32 distinct tunable capacitance statesbetween the RF+ terminal 780 and the RF− terminal 718 (e.g., numberedsuch that the states range between a tuning state 0 through a tuningstate 31). The values of each of the equivalent resistors and capacitorsused to implement the simplified equivalent circuit 792 are determinedin accordance with the mathematical formulae set forth in FIG. 7I,wherein each mathematical formula is shown in FIG. 7I adjacent to itscorresponding and associated component. For example, an equivalentR_(MIM)/k resistor 794 value is determined in accordance with themathematical formula shown adjacent to the resistor 794 in FIG. 7I(i.e., it is equal to the value of R_(MIM)/k, wherein k is selectedtunable state of the DTC 792 (or the “decimal equivalent” of the binarydigital control word applied to control the DTC 792). Similarly, thecapacitance value of an equivalent m*C_(MIM) capacitor 796 is determinedin accordance with the associated and corresponding mathematicalexpression shown adjacent to the capacitor 796 in FIG. 7I (i.e., it isdetermined in accordance with the mathematical expression of(m*C_(MIM)), wherein m=(2^(b)-1)-k; k is the selected tunable state ofthe DTC 792, and b is the number of DTC control bits. The values of theremaining equivalent circuit components are similarly determined inaccordance to their associated and corresponding mathematicalexpressions as set forth in FIG. 7I. The definition of the terms used inthe mathematical expressions of FIG. 7I are described above in detailwith reference to FIGS. 7A-7H. Exemplary values for some of these termsare also set forth in FIG. 7I. These exemplary values are identical tothose described above with reference to Equations 6a and 6b.

As described above with reference to the DTC implementations of FIGS.6A, 6B, and FIGS. 7A-7I, DTCs made in accordance with the presentteachings are implemented using an arrangement of stacked FETs. Thestacked FETs aid the present DTCs in meeting high power requirementsimposed by system standards. FIG. 7J shows how an “effective” FET stack“height” (i.e., the effective number of FETs in a FET stack) is achievedusing the present teachings, wherein the effective stack height exceedsthe actual stack height of the DTC (i.e., wherein the actual stackheight equals the number of FETs used in implementing the FET stack).The circuit 798 of FIG. 7J shows how the power handling of the DTC isincreased by using FET stacking and by positioning the MIM capacitor 799at the top of the FET stack 797. Due to the voltage division that occursbetween the MIM capacitor 799 and the FET stack 797 when the stackedFETs are turned OFF, the effective stack height (n_(eff)) is therebyincreased beyond the actual FET stack height. This further increases theDTC power handling as shown in the circuit of FIG. 7J. For this example,the stack height for FETs is 6, but the effective stack height due tothe MIM capacitor is 8.8.

As shown in FIG. 7J, the effective stack height (n_(eff)) may becalculated in accordance with the following mathematical expression:

${n_{eff} = {{n + \frac{C_{OFF}}{C_{MIM}}} = {{6 + \frac{0.28\mspace{14mu}{pF}}{0.1\mspace{14mu}{pF}}} = 8.8}}};$wherein n_(eff) comprises the effective stack height, n comprises thenumber of FETs in the stack, C_(MIM) comprises the capacitance value ofthe MIM capacitor 799, and C_(OFF) comprises the OFF capacitance of asingle FET of the FET stack 797 such as FET 795. As described above,given the exemplary values set forth in FIG. 7J, the effective stackheight is 8.8, while the actual stack height is 6.

Table 1 below shows exemplary design characteristics for an exemplary 1GHz DTC and 2 GHz DTC made in accordance with the present disclosure. Asshown in Table 1, in the embodiment shown, the exemplary 1 GHz DTC usesa 5 bit control word and six stacked FETs. The exemplary 2 GHz DTC usesa 5 bit control word and five stacked FETs. The total area occupied bythe exemplary 1 GHz DTC is 0.886 mm², and the total area occupied by theexemplary 2 GHz DTC is 0.402 mm².

TABLE 1 1 GHz DTC capacitor TR 25-Nov-07 Inputs: CMIM 0.25 [pF] COFF0.073333 [pF] Bits 5 Stack 6 RonCoff 0.6 [pF * ohm] Fet_area 0.0064[mm2], 1 ohm single fet MIM_area 0.0017 [mm2], 1 pF cap Vds_lin 2.54[Vpk] Vds_max 3.50 [Vpk] Outputs: Unit cells 31 Ron 8.18 Cmin 1.81 [pF]Cmax 8.00 [pF] Cstep 0.193 [pF] Range 4.41 Q900 86.5 Q1800 43.2 Q220035.4 FET area 0.873 [mm{circumflex over ( )}2] MIM area 0.013[mm{circumflex over ( )}2] Total area 0.886 [mm{circumflex over ( )}2]Vpk_lin 19.7 [Vpk] Vpk_max 27.2 [Vpk] Pwr_lin 35.9 [dBm] Pwr_max 38.7[dBm] 2 GHz DTC capacitor TR 25-Nov-07 Inputs: CMIM 0.1 [pF] COFF 0.048[pF] Bits 5 Stack 5 RonCoff 0.6 [pF * ohm] Fet_area 0.0064 [mm2], 1 ohmsingle fet MIM_area 0.0017 [mm2], 1 pF cap Vds_lin 2.54 [Vpk] Vds_max3.50 [Vpk] Outputs: Unit cells 31 Ron 12.5 Cmin 1.04 [pF] Cmax 3.20 [pF]Cstep 0.068 [pF] Range 3.08 Q900 141.5 Q1800 70.7 Q2200 57.9 FET area0.397 [mm{circumflex over ( )}2] MIM area 0.005 [mm{circumflex over( )}2] Total area 0.402 [mm{circumflex over ( )}2] Vpk_lin 18.8 [Vpk]Vpk_max 25.9 [Vpk] Pwr_lin 35.5 [dBm] Pwr_max 38.3 [dBm]

FIG. 8A shows a schematic of an exemplary 1 GHz DTC 800 made inaccordance with the design characteristics set forth in Table 1. Asshown in FIG. 8A, all of the significant bit sub-circuits are binaryweighted. As described above, both the MIM capacitors and the stackingFETs are binary weighted from a LSB to a MSB. The DTC is designed inaccordance with the above-described unit cell design approach. FIG. 8Ashows the total on resistance (Ron) and off capacitance (Coff) of eachof the significant bit sub-circuits (also referred to herein as circuitelements). For example, the LSB significant bit sub-circuit has an onresistance (Ron) of 8.18 ohms, and an off capacitance (Coff) of 0.0733pF. The next significant bit sub-circuit has an on resistance (Ron) of4.09 ohms, and an off capacitance (Coff) of 0.147 pF. The DTC 800 uses a5 bit control word which yields a “capacitance step” (Cstep) of 0.2 pF.The capacitance step comprises the difference in total capacitance ofthe DTC achieved by changing from a selected capacitance level (selectedcontrol word value) to the next significant bit capacitance level(selected control word value increased by a least significant bit [e.g.,by changing the control word from “00000” to “00001”]). The tuningrange, or tuning ratio, (defined herein as “C_(max)/C_(min)”) of the DTCis approximately 4.41 (i.e., the DTC achieves a total capacitance ofapproximately 1.81 pF when the control word is 00000, and it achieves atotal capacitance of approximately 8.0 pF when the control word is11111). As shown in FIG. 8A, the stacked FETs comprise six stacked FETs(i.e., n=6). T_(IN) equals 0.4 wherein T_(IN) comprises a “flavor” ofthe FET used in this exemplary embodiment. In the embodiment shown, inthis particular example a thick-oxide IN transistor with 0.4 μm gatelength is used. However, the skilled person will recognize other“flavors” of FETs can be used without departing from the spirit andscope of the present teachings. RonCoff of the DTC=600 fF-Ω. Theswitching time of the DTC is equal to R_(G)*C_(ON), which in the DTC 800of FIG. 8A equals 2.9 μs.

FIG. 8B shows a model simulation of the 1 GHz DTC 800 of FIG. 8A. Insome embodiments, for ESD protection reasons, a stack of FETs (i.e., astack-of-8) is inserted from RF pin to GND pin. This additional FETstack does not typically include a MIM capacitor on top of the FETstack, and it is always biased OFF. This stack of FETs protects the MIMcapacitors in the event of an ESD strike. The “ESD stack” is small andtypically smaller than is the unit cell (LSB bit). FIG. 8C shows a plot802 of the total capacitance of the DTC (y axis) versus the DTCcapacitance control word setting (i.e., it shows the total capacitanceof the DTC 800 as the 5-bit control word is increased from a minimumsetting of zero (binary 00000) to a maximum setting of 31 (binary11111)). As shown by the plot 802, advantageously, the total capacitanceof the DTC 800 increases linearly with respect to the capacitancecontrol setting. This is an improvement over the prior art tunablecapacitors which tended to have a non-linear plot of total capacitanceversus capacitance setting. The tuning range of the DTC comprises 1.79pF to 8.0 pF, and the Cstep is 0.194 pF.

FIG. 8D shows a plot 804 of the total Q-factor value of the DTC for agiven applied signal frequency (in this example, the Q-factor ismeasured at a signal frequency of 900 MHz) versus the DTC capacitancecontrol word setting (i.e., it shows the total Q-factor value of the DTC800 as the 5-bit control word is increased from a minimum setting ofzero (binary 00000) to a maximum setting of 31 (binary 11111)). As shownby the plot 804, advantageously, the total Q-factor value of the DTC 800remains relatively constant over the entire tuning range. For example,as shown in FIG. 8D, the total Q-factor of the DTC at a signal frequencyof 900 MHz remains at approximately 100 over all of the possiblecapacitance settings. This is a tremendous improvement over the priorart tunable capacitors which exhibited a non-linear relationship betweenthe total Q-factor values and capacitance settings. For example, theprior art tunable capacitor solutions exhibited Q-factor plots similarto the exemplary plots 806 and 808. As shown in FIG. 8D, the exemplaryprior art Q-factor plot 806 shows a non-linearly increasing Q-factorvalue with an increasing capacitance setting. The exemplary prior artQ-factor plot 808 also shows a non-linearly decreasing Q-factor valuewith an increasing capacitance setting.

FIGS. 8E and 8F show exemplary integrated circuit layout representationsof the 1 GHz DTC described above. More specifically, FIG. 8E shows anexemplary integrated circuit layout of a 1×bit LSB unit cell 820 made inaccordance with the present teachings. As shown in FIG. 8E, the 1×bitLSB unit cell 820 comprises a stack of 6 FETs (FETs 822, 824, 826, 828,830, and 832), wherein the FETs are coupled together in series, andwherein the stack of FETs is coupled in series to a 1×bit MIM capacitor834. FIG. 8E shows a close-up layout 840 of the topmost portion of the1×bit LSB unit cell 820. As shown in FIG. 8E, the close-up layout 840shows more details of the topmost FET (i.e., the FET 832) and the MIMcapacitor 834. Details of the gate resistors R_(G) (“bias” resistors)842 and drain-to-source R_(DS) resistors 844 are also shown in theclose-up layout 840 of FIG. 8E. In the embodiment shown in FIG. 8E, theMIM capacitor 834 comprises a 0.25 pF capacitor, the FETs have an ONresistance R_(ON) of 1.36 ohms, and an OFF capacitance C_(OFF) of 0.44pF.

FIG. 8F shows an exemplary integrated circuit layout 850 of the 1 GHzDTC described above. The capacitor control word (b₀, b₁, b₂, b₃, and b₄)852 is coupled to the DTC via electrostatic discharge protectioncircuitry 854. As shown in FIG. 8F, the 1×bit LSB unit cell comprises a1×bit LSB unit cell 820 of FIG. 8E. In the embodiment shown in FIG. 8F,each increasingly more significant bit sub-circuit is binary weighted asdescribed above. For instance, the second significant bit sub-circuit856 comprises 2*the LSB unit cell 820. The next significant bitsub-circuit (or element) 858 comprises 4*the LSB unit cell 820, and soon. The MIM capacitors are also binary weighted as described in moredetail above. For example, the second significant bit sub-circuit 856comprises a MIM capacitor 860 that is twice the size of the MIMcapacitor 834 of the LSB unit cell 820 (as shown in FIG. 8F, in someexemplary embodiments the 2× MIM capacitor 860 comprises twoinstantiations of the 1× LSB MIM capacitor 834). The next significantbit sub-circuit (or element) 858 comprises a MIM capacitor 862 that is4× the size of the MIM capacitor 834 of the LSB unit cell 820 (or it maycomprise four instantiations of the LSB MIM capacitor 834), and so on.

FIG. 9A shows a schematic of an exemplary 2 GHz DTC 900 made inaccordance with the design characteristics set forth in Table 1. Asshown in FIG. 9A, all of the significant bit sub-circuits are binaryweighted. As described above, both the MIM capacitors and the stackingFETs are binary weighted from a LSB to a MSB. The DTC is designed inaccordance with the above-described unit cell design approach. FIG. 9Ashows the total on resistance (Ron) and off capacitance (Coff) of eachof the significant bit sub-circuits (also referred to herein as circuitelements). For example, the LSB significant bit sub-circuit has an onresistance (Ron) of 12.5 ohms, and an off capacitance (Coff) of 0.048pF. The next significant bit sub-circuit has an on resistance (Ron) of6.25 ohms, and an off capacitance (Coff) of 0.096 pF. The DTC 900 uses a5 bit control word which yields a capacitance step (Cstep) of 0.067 pF.The tuning range, or tuning ratio, (defined herein as “C_(max)/C_(min)”)of the DTC is approximately 3.08 (i.e., the DTC achieves a totalcapacitance of approximately 1.04 pF when the control word is 00000, andit achieves a total capacitance of approximately 3.2 pF when the controlword is 11111). As shown in FIG. 9A, the stacked FETs comprise fivestacked FETs (i.e., n=5). T_(IN) equals 0.4 and RonCoff of the DTC=600fF-Ω. The switching time of the DTC is equal to R_(G)*C_(ON), which inthe DTC 900 of FIG. 9A equals 0.8 μs.

FIG. 9B shows a model simulation of the 2 GHz DTC 900 of FIG. 9A. FIG.9C shows a plot 902 of the total capacitance of the DTC (y axis) versusthe DTC capacitance control word setting (i.e., it shows the totalcapacitance of the DTC 800 as the 5-bit control word is increased from aminimum setting of zero (binary 00000) to a maximum setting of 31(binary 11111)). As shown by the plot 902, advantageously, the totalcapacitance of the DTC 900 increases linearly with respect to thecapacitance control setting. The tuning range of the DTC comprises 1.03pF to 3.20 pF, and the Cstep is 0.0678 pF.

FIG. 9D shows a plot 904 of the total Q-factor value of the DTC 900 fora given applied signal frequency (in this example, the Q-factor ismeasured at a signal frequency of 220 MHz) versus the DTC capacitancecontrol word setting. As shown by the plot 904, advantageously, thetotal Q-factor value of the DTC 900 remains relatively constant over theentire tuning range. FIGS. 9E and 9F show exemplary integrated circuitlayout representations of the 2 GHz DTC described above. Morespecifically, FIG. 9E shows an exemplary integrated circuit layout of a1×bit LSB unit cell 920 made in accordance with the present teachings.As shown in FIG. 9E, the 1×bit LSB unit cell 920 comprises a stack of 5FETs (FETs 922, 924, 926, 928, and 930), wherein the FETs are coupledtogether in series, and wherein the stack of FETs is also coupled inseries to a 1×bit MIM capacitor 934. FIG. 9E shows a close-up layout 940of the topmost portion of the 1×bit LSB unit cell 920. As shown in FIG.9E, the close-up layout 940 shows more details of the topmost FET (i.e.,the FET 930) and the MIM capacitor 934. Details of the gate resistorsR_(G) (“bias” resistors) 942 and drain-to-source R_(DS) resistors arealso shown in the close-up layout 940 of FIG. 9E. In the embodimentshown in FIG. 9E, the MIM capacitor 934 comprises a 0.1 pF MIMcapacitor, the FETs have an ON resistance R_(ON) of 2.5 ohms, and an OFFcapacitance C_(OFF) of 0.24 pF.

FIG. 9F shows an exemplary integrated circuit layout 950 of the 2 GHzDTC described above. The capacitor control word (b₀, b₁, b₂, b₃, and b₄)952 is coupled to the DTC via electrostatic discharge protectioncircuitry 954. As shown in FIG. 9F, the 1×bit LSB unit cell comprises a1×bit LSB unit cell 920 of FIG. 9E. In the embodiment shown in FIG. 9F,each increasingly more significant bit sub-circuit is binary weighted asdescribed above. For instance, the second significant bit sub-circuit956 comprises 2*the LSB unit cell 920. The next significant bitsub-circuit (or element) 958 comprises 4*the LSB unit cell 920, and soon. The MIM capacitors are also binary weighted as described in moredetail above. For example, the second significant bit sub-circuit 956comprises a MIM capacitor 960 that is twice the size of the MIMcapacitor 934 of the LSB unit cell 920 (as shown in FIG. 9F, in someexemplary embodiments the 2× MIM capacitor 960 comprises twoinstantiations of the 1× LSB MIM capacitor 934). The next significantbit sub-circuit (or element) 958 comprises a MIM capacitor 962 that is4× the size of the MIM capacitor 934 of the LSB unit cell 920 (or it maycomprise four instantiations of the 1× LSB MIM capacitor 934), and soon.

FIGS. 10A and 10B show a comparison of the capacitance tuning curves ofthe above-described DTCs with those of thin-film Barium StrontiumTitanate (BST) tunable capacitors. More specifically, FIG. 10A shows aplot 1000 of the DTC total capacitance versus the capacitance controlsetting for a DTC made in accordance with the present teachings. FIG.10B shows a plot 1002 of the BST capacitance versus the bias voltage fora BST tunable capacitor. As described above in more detail withreference to the plots of FIGS. 8C and 9C, and as shown in FIG. 10A,advantageously, the total capacitance of the DTC made in accordance withthe present teachings increases linearly with respect to the capacitancecontrol setting. This is an improvement over the prior art BST tunablecapacitors which have non-linear plots similar to the curve 1002 oftotal capacitance versus capacitance setting (i.e., bias voltage). Inaddition, the BST tunable capacitors also suffer from problemsassociated with hysteresis. The DTCs made in accordance with the presentteachings advantageously do not have this drawback.

DTC Design “Trade-Offs” and Design Considerations

The above-described digitally tuned capacitor (DTC) method and apparatusadvantageously may be designed to optimize or satisfy a wide range ofcircuit performance and DTC size characteristics. Using these designcharacteristics and “trade-offs”, the DTCs can be customized andoptimized to satisfy specifications and requirements imposed by systemproviders.

Design Trade-Offs—Tuning Range vs. Frequency at Q_(min) Values

For example, FIG. 11 shows a graph 1100 of the tuning range of a DTCversus the frequency of the applied signal for a selected minimumQ-factor value (Q_(min)). Two curves are shown in FIG. 11. A first curve1102 shows the tuning range of a first DTC having a minimum Q-factor of50 as the tuning range varies as a function of the frequency of theapplied signal. A second curve 1104 shows the tuning range of a secondDTC having a minimum Q-factor of 100 as the tuning range varies as afunction of the frequency of the applied signal. As described above withreference to Equations 1-6, the minimum Q-factor value (Q_(min)) and thetuning range (C_(max)/C_(min)) are strongly related to each other due tothe operating principle and design of the DTC of the present teachings.The minimum Q-factor of the present DTC is dependent on the product ofthe DTC ON resistance R_(ON) and the DTC OFF capacitance C_(OFF) (i.e.,R_(ON)C_(OFF)). The curves 1102 and 1104 of FIG. 11 are plotted assumingthat RonCoff=600 fF-Ω.

In some embodiments, the tuning range is determined in accordance withthe following Equation 7:

$\begin{matrix}{{Tuning\_ range} = {\frac{1}{{\omega \cdot R_{ON}}{C_{OFF} \cdot Q_{\min}}} + 1.}} & {{Equation}\mspace{14mu} 7}\end{matrix}$

Equation 7 shows the limitation of the tuning range based on the RonCoffof the process and Q requirement. Equation 3 teaches how to choose theratio between Coff and CMIM based on the required Tuning ratiospecification. The tuning range “rule-of-thumb” design characteristicsare set forth in Table 2 below:

TABLE 2 Frequency of Applied Signal Q = 50 Q = 100  900 MHz Tuning Ratio= 7:1 Tuning Ratio = 4:1 1800 MHz Tuning Ratio = 4:1 Tuning Ratio =2.5:1 2200 MHz Tuning Ratio = 3.4:1 Tuning Ratio = 2.2:1Design Trade-Off—Tuning Range and Die Area vs. Q_(min) Values

As described above with reference to the DTCs of FIGS. 4A, 5A-5B, 6, 8A,8F, 9A, and 9F, owing to the operating principles and design techniquesof the present DTC teachings, the die area occupied by the DTC increasesand its associated tuning range decreases with increasing Q-factorvalues. This design trade-off is shown diagrammatically in the graph1200 of FIG. 12. FIG. 12 shows a graph 1200 of the tuning range and diearea requirements versus minimum Q-factor values for a selected DTC at agiven applied signal frequency (in the plot 1200 of FIG. 12 the curves1202 and 1204 are plotted assuming that the applied signal frequencycomprises 900 MHz). Referring again to FIG. 12, the curve 1202 shows howthe tuning range decreases as the Q-factor requirement of the DTCincreases. The curve 1204 shows how the die area requirement of the DTCincreases as the Q-factor requirement of the DTC increases. The curves1202 and 1204 of FIG. 12 are plotted assuming the following: RonCoff=600fF-Ω; the frequency of the applied signal is 900 MHz; the capacitancecontrol word is 5 bits; a FET Stack-of-6 (i.e., n=6); and Cmax=8.2 pF.

Design Trade-Off—Cmax vs. FET Die Area

As described above with reference to the DTCs of FIGS. 4A, 5A-5B, 6, 8A,8F, 9A, and 9F, owing to the operating principles and design techniquesof the present DTC teachings, the die area requirements of the DTC isproportional to its maximum capacitance (Cmax) (i.e., the maximumcapacitance achievable by the DTC). Referring now to FIG. 13, curve 1302shows how the FET die area requirement (i.e., the die area requirementof the FETs of the DTC) increases as the maximum DTC capacitance (Cmax)increases. Curve 1302 shows the relationship between the FET die areaand Cmax for a DTC having a stack of six shunting FETs (therebyproviding a DTC power handling capability of approximately +35 dBM).Curve 1304 shows the relationship between the FET die area and Cmax fora DTC having a stack of five shunting FETs (thereby providing a DTCpower handling capability of approximately +33.4 dBM). As expected, theDTC having the lower power handling requirement (and fewer number ofstacked FETs) occupies less die area than does the DTC having the higherpower handling requirement (and therefore requiring a greater number ofstacked FETs). The plots 1302, 1304 assume the following: Q=50 @ 900MHz; a capacitance control word of 5 bits, and that the DTC has a 7:1tuning range. It is also assumed that the FET (IN 0.4) area comprises 80um×80 um for a 1Ω single FET.

Design Considerations—Optimizing DTC for Reduced IC Die Area

Reducing the Cmax of the DTC—Several design trade-offs and designconsiderations can be taken advantage of in order to reduce theintegrated circuit die area occupied by the DTC. For example, asdescribed above with reference to FIG. 13, owing to the operatingprinciples and design techniques of the present DTC teachings, the diearea requirements of the DTC is proportional to its maximum capacitance(Cmax) (i.e., the maximum capacitance achievable by the DTC). Therefore,if the maximum capacitance of the DTC (Cmax) can be reduced, the diearea required by the DTC can also be reduced. Consequently, in order tooptimize the DTC for the smallest possible die area, it is useful to usea tuner topology that requires the smallest maximum capacitance Cmax.For example, a coupled-resonator tuner topology is significantly better.As an example, if the maximum capacitance Cmax specification of aselected DTC is reduced from 9.4 pF to 6.0 pF, the die area required bythe selected DTC made in accordance with the present teachings isreduced by 36%=[(1)-(6/9.4)].

Reducing the FET Stack Height of the DTC—As described above withreference to FIG. 13, if the power handling requirements of the DTC arereduced, the DTC can be implemented using a fewer number n of stackedFETs, and the die area required by the DTC can also thereby be reduced.As described in more detail above, the power handling requirementsimposed upon the DTC dictate the number n of stacked FETs required toimplement the unit cell sub-circuits. Therefore, if the power handlingspecification can be relaxed, the integrated circuit die area requiredby the DTC can be reduced. For example, a reduction of the maximum powerhandling specification from +38.5 dBM to +36.6 dBM results in a reduceddie area of approximately 30% for a selected DTC made in accordance withthe present teachings. If the DTCs are implemented in theabove-described UltraCMOS™ process technology, the DTCs are notsensitive to power. However, the UltraCMOS DTC is sensitive to thevoltage swing of the applied RF signal which can theoretically double ininfinite mismatch. However, when the DTC is used in a mobile handset,for example, the power amplifiers of the handset do not typicallyproduce very high voltages. Therefore, the power handling specificationof the DTC may be able to be relaxed, resulting in reduced powerhandling requirements, a lower number n of stacked FETs (i.e., a lowerstacked FET height), and a correspondingly reduced IC die area.

Placing a Fixed MIM Capacitor in Parallel with the DTC—As describedabove with reference to the DTCs of FIGS. 4A, 5A-5B, 6, 8A, 8F, 9A, and9F, owing to the operating principles and design techniques of thepresent DTC teachings, the tuning ratio of the DTC (Cmax/Cmin) isdetermined by the DTC Q-factor value imposed by system specifications,the frequency of the applied signal, and the R_(ON)C_(OFF)figure-of-merit of the switch process (i.e., the product of the DTC ONresistance R_(ON) and the DTC OFF capacitance C_(OFF)). If the tuningratio for a selected DTC is greater than a tuning ratio required by asystem specification, a fixed MIM capacitor can be added (C_(ADD)), orcoupled in parallel, to the DTC resulting in a significantly scaled-downversion of the selected DTC. The scaled-down DTC occupies much less ICdie area than its unmodified DTC counterpart, yet it still meets all ofthe required system specifications (such as, for example, tuning ratio,Q-factor minimum value, Cmax, etc.).

In one embodiment of a DTC modified to include an added capacitorC_(ADD) coupled in parallel to the unmodified DTC, and assuming theQ-factor remains the same, changing the tuning ratio of the modified DTC(i.e., the tuning ratio of the circuit comprising the unmodified DTCcoupled in parallel with the fixed MIM capacitor C_(ADD)) from 4.7:1 to3:1 reduces the DTC die area by approximately 30%=[1−3/4.7]. Relaxing orreducing the Q-factor requirements of the DTC results in increasedtuning ratios. This leads to even greater IC die area reduction when afixed MIM capacitor C_(ADD) is coupled in parallel to the DTC. Reducingthe Q-factor value from 80 to 60 increases the DTC tuning ratio from4.7:1 to 5.9:1. If the tuning ratio of the modified DTC is then forcedto be 3:1, for example, with a fixed capacitor coupled in parallel tothe DTC, this reduces the DTC die area by approximately62%=[1−60/80*3/5.9].

Table 3 set forth below shows the reduction in die area occupied by agiven DTC that is achieved by taking advantage of the designconsiderations and trade-offs described above. As shown in Table 3, forthe given reductions in Cmax (9.4 pF to 6.0 pF), Tuning Ratio (4.7:1 to3:1), Q-factor (80 to 60), linear power (35.7 dBm to 33.8 dBm), and MaxPower (38.5 dBM to 36.6 dBM), and by modifying the DTC to include afixed MIM capacitor C_(ADD) coupled in parallel to the DTC, anapproximately 70% IC reduction in die area required to implement the DTCcan be realized in accordance with the present disclosure (i.e., from0.96 mm² to 0.29 mm²).

TABLE 3 Design A Design B Max cap (Cmax)  9.4 pF  6.0 pF Min cap (Cmin) 2.0 pF  2.0 pF Tuning Ratio 4.7:1 3:1 Q @ 900 MHz 80 60 Linear Power35.7 dBm 33.8 dBm Max Power 38.5 dBm 36.6 dBm Die Area 0.96 mm² 0.29 mm²Tuning Range and Die Area vs. Q_(min) Values for Unmodified and ModifiedDTCs

As noted above, the IC die area required by a DTC can be reduced incases where the DTC tuning ratio exceeds that imposed by systemspecifications. This reduction can be achieved by coupling a fixed MIMcapacitor (C_(ADD)) in parallel with the DTC. FIGS. 14A and 14B showgraphs of the tuning ranges and die area requirements versus minimumQ-factor values for a selected unmodified DTC (graph 1400 of FIG. 14A)and a modified DTC (graph 1400′ of FIG. 14B, wherein the selected DTC ismodified by coupling a fixed MIM capacitor C_(ADD) in parallel to theDTC) at a given applied signal frequency. In the example shown, thecurves of the graphs 1400, 1400′ are plotted assuming that the appliedsignal frequency comprises 900 MHz. The graph 1400 is identical to thegraph 1200 described above with reference to FIG. 12. Referring now toFIG. 14A, the curve 1402 shows how the tuning range decreases as theQ-factor requirement of the DTC increases. The curve 1404 shows how thedie area requirement of the DTC increases as the Q-factor requirement ofthe DTC increases. The curves 1402 and 1404 of FIG. 14A are plottedassuming the following conditions: RonCoff=600 fF-Ω; the frequency ofthe applied signal is 900 MHz; the capacitance control word is 5 bits; aFET Stack-of-6 (i.e., n=6); and Cmax=8.2 pF. As shown in FIG. 14A, andspecifically as shown by the curve 1404, for a given unmodified DTChaving a Q-factor value of 80, a capacitance range of 1.7 to 8.0 pF, andtherefore a tuning range (or tuning ratio) of 4.7:1, the die arearequired for the given unmodified DTC equals 0.82 mm².

As noted above, if the tuning range of a given unmodified DTC exceedsthat required by system specifications, the DTC can be modified bycoupling a fixed MIM capacitor (C_(ADD)) in parallel to the unmodifiedDTC, resulting in a reduction of the IC die area occupied by themodified DTC. The graph 1400′ of FIG. 14B diagrammatically shows thisphenomenon. The curve 1402′ shows that the tuning range of the modifiedDTC can be forced to be relatively stable (i.e., it does not varysignificantly) as the Q-factor requirement of the modified DTCincreases. The curve 1404′, similar to the curve 1404 of FIG. 14A, showshow the die area requirement of the modified DTC increases as theQ-factor requirement of the DTC increases. The curves 1402′ and 1404′ ofFIG. 14B are plotted assuming the following conditions: RonCoff=600fF-Ω; the frequency of the applied signal is 900 MHz; the capacitancecontrol word is 5 bits; a FET Stack-of-6 (i.e., n=6); and Cmax=8.2 pF.As shown in FIG. 14B, assuming a Q-factor of 80, for example, andspecifically as shown by the curve 1402′, the tuning range of themodified DTC is reduced from 4.7:1 (see curve 1404) (i.e., C rangingbetween 1.7 and 8.0 pF) to 3:1 (see curve 1404′) (i.e., C rangingbetween 2.67 and 8.0 pF).

As shown in FIG. 14B and specifically as shown by the curve 1404′, for agiven modified DTC having a Q-factor value of 80, a capacitance range of2.67 to 8.0 pF, and therefore a tuning range (or tuning ratio) of 3:1,the die area required for the modified DTC equals 0.54 mm². Thus, byreducing the tuning range (i.e., forcing the tuning range to be lowedthan its unmodified counterpart DTC) of the modified DTC, and modifyingthe DTC to include a fixed MIM capacitor coupled in parallel to the DTC,a reduction of approximately 34% (1−0.54 mm²/0.82 mm²) in IC die area isachieved for a Q-factor of 80. As shown by the curve 1404′ of FIG. 14B,different die area savings are achievable using this same designtechnique for different Q-factor values.

Idealized Equations Governing C_(ADD) and the Design Parameters of theModified DTC—As described above with reference to FIGS. 14A and 14B, incases wherein a selected DTC has a tuning ratio exceeding that requiredby the system specifications (such as those imposed by the variouswireless telecommunication standards), the selected DTC can be modifiedwith a fixed MIM capacitor C_(ADD) coupled in parallel to the DTC. For agiven minimum Q-factor of the DTC, such a modification reduces the ICdie area otherwise required by the DTC while also maintaining all othersystem performance requirements. The fixed MIM capacitor C_(ADD)comprises an “ideal” capacitor because it is independent of the ONresistance R_(ON) of the DTC. In contrast to the MIM capacitors of theDTC sub-circuits (e.g., the 1×LSB MIM capacitor of the unit cell block),the fixed MIM capacitor C_(ADD) is not switched by a switching FET.Rather, C_(ADD) is constantly applied between the terminals of the DTC).C_(ADD) increases the total effective Q-factor value of the combinedcircuit comprising the unmodified (or “original”) DTC coupled to thefixed MIM capacitor C_(ADD). Consequently, the Q-factor value of the DTCadvantageously can be reduced because C_(ADD) helps to keep the totalQ-factor of the combined circuit at the value required by the systemspecifications. The DTC therefore is re-designed to have a loweredQ-factor value. To compensate for the other effects that the additionalfixed capacitance of C_(ADD) has on the combined DTC-C_(ADD) circuit,the DTC is also re-designed to have a lower maximum total capacitance(Cmax) and higher tuning ratio (TR). In one embodiment, the parasiticcapacitance within the DTC should be lumped together with thecapacitance of C_(ADD).

As shown in FIG. 15A, the unmodified DTC has a minimum total capacitance(Cmin) of 1.65 pF; a maximum total capacitance (Cmax) of 7.75 pF;thereby producing a Tuning Ratio (or Tuning Range, TR) of 4.70:1 [asTR=(Cmax/Cmin):1, by definition]; and a minimum Q-factor value Qmin of80. FIG. 15B shows how the DTC 1500 of FIG. 15A is modified with C_(ADD)to produce the modified DTC 1500′. FIG. 15B also shows idealizedequations (described in more detail below) that are used to re-design(i.e., modify) the DTC 1500 to produce a modified DTC 1501 having areduced Q-factor value, a lower maximum total capacitance (Cmax) and ahigher tuning ratio (TR). FIG. 15B also shows the idealized equation fordetermining the value of C_(ADD).

As shown in FIG. 15B, in one embodiment, the modified DTC 1500′ isdesigned in accordance with the following equations:

$\begin{matrix}{{C_{ADD} = \frac{C_{\max}^{2} - {C_{\min}C_{\max}{TR}}}{C_{\min} + {\left( {C_{\max} - {2C_{\min}}} \right){TR}}}};} & {{Equation}\mspace{14mu} 8} \\{{C_{\min,2} = {\frac{C_{\max}}{TR} - C_{ADD}}};} & {{Equation}\mspace{14mu} 9} \\{{C_{\max,2} = {C_{\max} - C_{ADD}}};} & {{Equation}\mspace{14mu} 10} \\{{Q_{\min,2} = {\frac{C_{\max} - C_{ADD}}{C_{\max}}Q_{\min}}};} & {{Equation}\mspace{14mu} 11}\end{matrix}$

wherein C_(ADD) comprises the capacitance of the fixed capacitor coupledin parallel to the modified DTC 1501; C_(min) comprises the minimumtotal capacitance of the unmodified DTC (i.e., the DTC 1500 of FIG.15A); C_(max) comprises the maximum total capacitance of the unmodifiedDTC 1500; Q_(min) comprises the minimum allowable Q-factor value of theunmodified DTC 1500, TR comprises the total Tuning Ratio (or TuningRange) of the re-designed and modified DTC 1500′ (i.e., the total TuningRange of the combined DTC-C_(ADD) circuit 1500′); C_(min,2) comprisesthe minimum total capacitance of the modified DTC 1501; C_(max,2)comprises the maximum total capacitance of the modified DTC 1501; andwherein Q_(min,2) comprises the minimum allowable Q-factor value of themodified DTC 1501. Note that the minimum allowable Q-factorQ_(MIN-total) of the entire modified DTC 1500′ (i.e., the Q_(MIN) of thecombined DTC-C_(ADD) circuit 1500′) is determined in accordance withEquation 12 set forth below:Q _(MIN-total) =Q _(min,2)/((C _(max) −C _(ADD))/C _(max)).  Equation12:

Using the idealized equations (Equations 8-11) set forth above, the DTCcircuit designer can readily design the modified DTC 1501 to have alower Q-factor value (i.e., a lowered minimum Q-factor value Q_(min,2)of the modified DTC 1501 is computed in accordance with Equation 11),and a lower maximum total capacitance (Cmax) (i.e., a lower maximumtotal capacitance C_(max,2) of the modified DTC 1501 is computed inaccordance with Equation 10). Equation 8 is used to calculate thecapacitance value of C_(ADD). The minimum total capacitance of themodified DTC 1501 C_(min,2) is computed in accordance with Equation 9.The tuning ratio of the modified DTC circuit alone 1501 (i.e., thetuning ratio of the DTC uncoupled from the C_(ADD)) is increased ascompared to the tuning ratio of the unmodified DTC 1500. However, thetuning ratio TR of the combined DTC-C_(ADD) circuit 1500′ can be forcedto be a lower tuning ratio (as compared with the TR of the unmodifiedDTC 1500). For example, as shown in FIGS. 15A and 15B, the TR of the DTCis lowered from 4.70:1 to 3:1. The value of TR for the combinedDTC-C_(ADD) circuit 1500′ is forced to 3:1 in this example.

As shown in FIGS. 15A and 15B, an exemplary unmodified DTC 1500 has thefollowing parameters: Cmin=1.65 pF; Cmax=7.75 pF, the tuning ratio istherefore equal to 4.70:1; and the DTC 500 has a minimum allowableQ-factor value of 80. Based on these DTC parameters, and using the abovedescribed Equations 8-11, the C_(ADD), C_(min,2), C_(max,2), andQ_(min,2) parameters of the modified DTC 1501 are computed. The TR ofthe DTC 1500′ is forced to be 3:1 in this example. The resultingexemplary calculations are set forth in the equations below:

$\begin{matrix}{{C_{ADD} = {\frac{C_{\max}^{2} - {C_{\min}C_{\max}{TR}}}{C_{\min} + {\left( {C_{\max} - {2C_{\min}}} \right){TR}}} = {\frac{7.75^{2} - {1.65 \cdot 7.75 \cdot 3}}{1.65 + {\left( {7.75 - {2 \cdot 1.65}} \right) \cdot 3}} = {1.45\mspace{14mu}{pF}}}}};} & {{Equation}\mspace{14mu} 8^{\prime}} \\{\mspace{79mu}{{C_{\min,2} = {{\frac{C_{\max}}{TR} - C_{ADD}} = {{\frac{7.75}{3} - 1.45} = {1.13\mspace{14mu}{pF}}}}};}} & {{Equation}\mspace{14mu} 9^{\prime}} \\{\mspace{79mu}{{C_{\max,2} = {{C_{\max} - C_{ADD}} = {{7.75 - 1.45} = {6.3\mspace{14mu}{pF}}}}};}} & {{Equation}\mspace{14mu} 10^{\prime}} \\{Q_{\min,2} = {{\frac{C_{\max} - C_{ADD}}{C_{\max}}Q_{\min}} = {{\frac{7.75 - 1.45}{7.75} \cdot 80} = {65.0.}}}} & {{Equation}\mspace{14mu} 11^{\prime}} \\{Q_{{MIN} - {total}} = {{Q_{\min,2}/\left( {\left( {C_{\max} - C_{ADD}} \right)/C_{\max}} \right)} = {65.0/\left( {{\left( {7.75 - 1.45} \right)/7.75} = {80.0.}} \right.}}} & {{Equation}\mspace{14mu} 12^{\prime}}\end{matrix}$

Therefore, in cases wherein the resulting DTC tuning ratio (based uponthe minimum allowable Q-factor value imposed by system specifications)is higher than that required by the specifications, a modified DTC 1501can be designed using a fixed MIM capacitor (C_(ADD)) coupled inparallel to the DTC 1501. The entire modified DTC 1500′ (i.e., combinedDTC 1501 and C_(ADD) capacitor circuit) meets the necessary systemspecifications but advantageously occupies less IC die area. In theexample given above and shown in FIGS. 15A and 15B, C_(ADD)=1.45 pF;Cmin,2=1.13 pF; Cmax,2=6.3 pF; and Qmin,2=65.0. The die area required toimplement the DTC is reduced from 0.82 mm² to 0.55 mm² (or byapproximately 33%). This aspect of the present teachings allows the DTCto be tailored to efficiently meet the design specifications andrequirements imposed by system standards. By taking advantage of thedesign trade-offs and considerations described above, precious die areasavings can be achieved yet still allow the present DTC to meet therequirements imposed by system specifications and standards.

While the FETs described above with reference to the present DTC methodand apparatus may comprise any convenient MOSFET device, in someembodiments they are implemented in accordance with improved process andintegrated circuit design advancements developed by the assignee of thepresent application. One such advancement comprises the so-called“HaRP™” technology enhancements developed by the assignee of the presentapplication. The HaRP enhancements provide for new RF architectures andimproved linearity in RF front end solutions. FETs made in accordancewith the HaRP enhancements are described in pending applications ownedby the assignee of the present application. For example, FETs made inaccordance with the HaRP enhancements are described in U.S. patent Ser.No. 11/484,370, filed Jul. 10, 2006, entitled “Method and Apparatus foruse in Improving Linearity of MOSFETs Using an Accumulated Charge Sink”;and in U.S. patent Ser. No. 11/520,912, filed Sep. 14, 2006, andentitled “Method and Apparatus Improving Gate Oxide Reliability byControlling Accumulated Charge”. Both of the above-cited patentapplications (i.e., application Ser. No. 11/484,370, filed Jul. 10, 2006and application Ser. No. 11/520,912, filed Sep. 14, 2006, areincorporated herein by reference as if set forth in full. As notedabove, in some embodiments, the FETs described above with reference tothe present DTC method and apparatus are implemented in accordance tothe teachings of these incorporated applications (application Ser. Nos.11/484,370 and 11/520,912).

More specifically, and as described in application Ser. No. 11/484,370,FETs made in accordance with HaRP technology enhancements compriseAccumulated Charge Control (ACC) SOI MOSFETs, wherein each ACC SOIMOSFET includes an Accumulated Charge Sink (ACS) coupled thereto whichis used to remove accumulated charge from the ACC FET body when the FEToperates in an accumulated charge regime. The ACS facilitates removal orotherwise controls the accumulated charge only when the ACC SOI MOSFEToperates in the accumulated charge regime. Thus, the HaRP technologyenhancements provide a method and apparatus for use in improvinglinearity characteristics of MOSFET devices using the accumulated chargesink (ACS). Via the ACS terminal, the HaRP FETs are adapted to remove,reduce, or otherwise control accumulated charge in SOI MOSFETs, therebyyielding improvements in FET performance characteristics. In oneexemplary embodiment, a circuit having at least one SOI MOSFET isconfigured to operate in an accumulated charge regime. The ACS isoperatively coupled to the body of the SOI MOSFET, and eliminates,removes or otherwise controls accumulated charge when the FET isoperated in the accumulated charge regime, thereby reducing thenonlinearity of the parasitic off-state source-to-drain capacitance ofthe SOI MOSFET. In RF switch circuits implemented with the improved SOIMOSFET devices, harmonic and intermodulation distortion is reduced byremoving or otherwise controlling the accumulated charge when the SOIMOSFET operates in an accumulated charge regime.

As described in the co-pending and above-incorporated application Ser.No. 11/484,370 patent application, in some embodiments the ACC MOSFETcomprises as a four terminal device, wherein an accumulated charge sink(ACS) terminal is coupled to a gate terminal via a diode. One such fourterminal ACC MOSFET 1503 is shown in FIG. 15C. FIG. 15C is a simplifiedschematic of an improved SOI NMOSFET 1503 adapted to control accumulatedcharge, embodied as a four terminal device, wherein the ACC MOSFET 1503includes a gate terminal 1502, source terminal 1504, drain terminal 1506and accumulated charge sink (ACS) terminal 1508. As shown in theembodiment of FIG. 15C, the ACS terminal 1508 is coupled to the gateterminal 1502 via a diode 1510. This embodiment may be used to prevent apositive current flow into the body of the MOSFET 1503 caused by apositive Vg-to-Vs (or, equivalently, Vgs, where Vgs=Vg−Vs) bias voltage,as may occur, for example, when the ACC MOSFET 1503 is biased into anon-state condition. When biased off, the ACS terminal voltage VACScomprises the gate voltage plus a voltage drop across the diode 1510. Atvery low ACS terminal current levels, the voltage drop across the diode1510 typically also is very low (e.g., <<500 mV, for example, for atypical threshold diode). The voltage drop across the diode 1510 may bereduced to approximately zero by using other diodes, such as a 0Vfdiode, for example. In one embodiment, reducing the voltage drop acrossthe diode is achieved by increasing the diode 1510 width. Additionally,maintaining the ACS-to-source or ACS-to-drain voltage (whichever biasvoltage of the two bias voltages is lower) increasingly negative, alsoimproves the linearity of the ACC MOSFET device 1503.

In some embodiments, as described above with reference to FIGS. 6A-6B,when the FETs are turned ON, a typical value of +2.75V voltage issupplied to their gate terminals. The FETs are turned OFF by applying atypically negative voltage of −3.4V. Supplying a larger level ofnegative voltage improves the linearity and harmonics performancecharacteristics of the FETs. Typically the negative voltage applied tothe FETs ranges between −1 and −3.6V. In one exemplary embodiment of thepresent DTC teachings, a negative voltage of −3.4V is applied.

For this reason in other embodiments of the present DTC teachings, asnoted above with regard to FIGS. 6A and 6B, a negative voltage generatoris included in an integrated circuit implementation of the DTC. Thenegative voltage generator is typically implemented as a charge pump.The charge pump provides, in one exemplary embodiment, the −3.4V voltagefrom the +2.75V supply voltage. In addition to the negative voltagegenerator, level shifters can be used to convert external controlsignals (e.g., between 0 and +2.75V) to −3.4V/+2.75v. The externalcontrol signals can be used to bias the FETs. In addition to thenegative voltage generator and the level shifters, this embodiment canalso include other blocks that provide additional support circuitry forthe DTC. For example, these other blocks may include serial bus, controlalgorithms, impedance mismatch detection circuitry, among otherfunctions.

The DTCs described above, and specifically the various significant bitsub-circuits (such as, for example, the LSB sub-circuit 602 of FIG. 6Bwhich comprises the unit cell design block) are described above ascomprising at least a plurality of stacked FETs coupled in series withcapacitors (in most of the embodiments described above, the capacitorscomprise MIM capacitors). While many applications may require orencourage implementation of the stacked switches using FETs, the presentDTC teachings also contemplate use of other switching devices toimplement the switching devices in series with the capacitors. Forexample, in some embodiments, the switching devices comprise laterallydiffused metal oxide semiconductor (LDMOS) transistors. In otherembodiments Micro-Electro-Mechanical Systems (MEMS) switches are used toimplement the switching devices. Further, as noted above, although mostof the DTCs described above implement the capacitors of the unit celldesign blocks with MIM capacitors, the present DTC is not so limited. Inother embodiments, the capacitors are implemented using other types ofcapacitance devices.

Conclusion

Availability of specification compliant tunable components will have asignificant impact on RF architectures for multi-band multi-modecellular phones. The present DTC methods and apparatus can be used inmany different environments and applications, including, but not limitedto adaptive impedance matching, antenna band and impedance tuning, PowerAmplifier (PA) output match tuning, RF filter and duplexer tuning,tunable and reconfigurable filters, antennas and PAs. The specificationhave difficult and difficult to meet requirements—high power handling(+35 dBm), high linearity (IMD3 −105 dBm), low-loss (Q>50-100), highreliability, 3:1-8:1 tuning range, fast switching speed (5 uS),inexpensive, mass-producible. The general requirements for tunablecomponents are very similar to the requirements for handset antennaswitches, which makes UltraCMOS implemented DTCs an excellent candidatetechnology to implement the DTCs described above. This particularimplementation relies heavily on the unique capability of stackingtransistors for high power handling and linearity and being able tointegrate high-Q capacitors. The UltraCMOS approach appears to be theonly monolithically integrated single-die solid-state tunable capacitorin existence that meets all the specifications, with all the samebenefits than UltraCMOS handset antenna switches. The DTCs describedabove advantageously can be produced in mass, at low-cost withhigh-reliability on a fully integrated device that is an alternative toMEMS and BST implementations. Proven high volume UltraCMOS switchtechnology can be used to implement the DTCs. This process technologyallows for monolithic integration of serial or parallel bus, digitalmismatch sensors, control algorithms that can also be used to supportthe present DTCs in some fully integrated solution embodiments.Advantageously, the DTC can be usable in impedance tuner applications,in antenna tuning, PA output match tuning, and many other usefulapplications.

A number of embodiments of the present invention have been described.Nevertheless, it will be understood that various modifications may bemade without departing from the spirit and scope of the claimedinvention.

Accordingly, it is to be understood that the invention is not to belimited by the specific illustrated embodiment, but only by the scope ofthe appended claims.

What is claimed is:
 1. A module comprising at least one integratedcircuit chip, wherein i) the at least one integrated circuit chip isincorporated into the module and includes: a first node; a second node;and a series arrangement of one or more capacitive elements and two ormore stacked switches coupled between the first node and the secondnode; ii) the two or more stacked switches are configured to withstand avoltage greater than a voltage withstood by one switch; iii) each of thetwo or more stacked switches has a control node connectable to a controlsignal via a resistive element; iv) the control signal is configured toenable or disable the two or more stacked switches, thereby adjustingthe capacitance between the first node and the second node, v) the twoor more stacked switches are stacked SOI MOSFET switches, and vi) theone or more capacitive elements have capacitance values assignedaccording to a capacitance weighting scheme and the two or more stackedswitches have switch sizes assigned according to a switch size weightingscheme, the switch size weighting scheme being a function of thecapacitance weighting scheme.
 2. The module of claim 1, wherein acapacitive element of the one or more capacitive elements connects a topswitch of the two or more stacked switches to the first node, the topswitch being the closest switch to the first node and the farthestswitch from the second node.
 3. The module of claim 1, wherein the twoor more stacked SOI MOSFET switches are consecutively stacked one oneach other without capacitive elements between the SOI MOSFET switchesand all of the one or more capacitive elements are arranged above a topswitch of the SOI MOSFET switches between the top switch and the firstnode.
 4. The module of claim 1, wherein a capacitive element of the oneor more capacitive elements connects a drain or source of a top SOIMOSFET switch of the two or more stacked SOI MOSFET switches to thefirst node.
 5. The module of claim 2, wherein the capacitive elementcomprises one or more capacitors.
 6. The module of claim 1, wherein theswitch size weighting scheme is same as the capacitance weightingscheme.
 7. The module of claim 2, wherein the second node is coupled toground.
 8. A communication device comprising: at least one integratedcircuit chip; wherein i) the at least one integrated circuit chip isincorporated into the communication device and further includes: a firstnode; a second node; and a series arrangement of one or more capacitiveelements and two or more stacked switches coupled between the first nodeand the second node; ii) the two or more stacked switches are configuredto withstand a voltage greater than a voltage withstood by one switch;iii) each of the two or more stacked switches has a control nodeconnected to a control signal via a resistive element, iv) the controlsignal is configured to enable or disable the two or more stackedswitches and thereby adjusting the capacitance between the first nodeand the second node, v) the two or more stacked switches are stacked SOIMOSFET switches, and vi) the one or more capacitive elements havecapacitance values assigned according to a capacitance weighting schemeand the two or more stacked switches have switch sizes assignedaccording to a switch size weighting scheme, the switch size weightingscheme being a function of the capacitance weighting scheme.
 9. Thecommunication device of claim 8, wherein a capacitive element of the oneor more capacitive elements connects a top switch of the two or morestacked switches to the first node, the top switch being the closestswitch to the first node and the farthest switch from the second node.10. The communication device of claim 8, wherein the two or more stackedSOI MOSFET switches are consecutively stacked one on each other withoutcapacitive elements between the SOI MOSFET switches and all of the oneor more capacitive elements are arranged above a top switch of the SOIMOSFET switches between the top switch and the first node.
 11. Thecommunication device of claim 8, wherein a capacitive element of the oneor more capacitive elements connects a drain or source of a top SOIMOSFET switch of the two or more stacked SOI MOSFET switches to thefirst node.
 12. The communication device of claim 9, wherein thecapacitive element comprises one or more capacitors.
 13. Thecommunication device of claim 8, wherein the switch size weightingscheme is same as the capacitance weighting scheme.
 14. An RF front-endcomprising: at least one integrated circuit chip and other RF front-endcomponents; wherein: i) the at least one integrated circuit chip andother RF front-end components are incorporated into the RF front-end,the at least one integrated circuit chip further includes: a first node;a second node; a series arrangement of one or more capacitive elementsand two or more stacked switches coupled between the first node and thesecond node; ii) the two or more stacked switches are configured towithstand a voltage greater than a voltage withstood by one switch; iii)each of the two or more stacked switches has a control node connected toa control signal via a resistive element, iv) the control signal isconfigured to enable or disable the two or more stacked switches andthereby adjusting the capacitance between the first node and the secondnode, v) the two or more stacked switches are stacked SOI MOSFETswitches, and vi) the one or more capacitive elements have capacitancevalues assigned according to a capacitance weighting scheme and the twoor more stacked switches have switch sizes assigned according to aswitch size weighting scheme, the switch size weighting scheme being afunction of the capacitance weighting scheme.
 15. The RF front-end ofclaim 14, wherein a capacitive element of the one or more capacitiveelements connects a top switch of the two or more stacked switches tothe first node, the top switch being the closest switch to the firstnode and the farthest switch from the second node.
 16. The RF front-endof claim 15, wherein the two or more stacked SOI MOSFET switches areconsecutively stacked one on each other without capacitive elementsbetween the SOI MOSFET switches and all of the one or more capacitiveelements are arranged above a top switch of the SOI MOSFET switchesbetween the top switch and the first node.
 17. The RF front-end of claim14, wherein a capacitive element of the one or more capacitive elementsconnects a drain or source of a top SOI MOSFET switch of the two or morestacked SOI MOSFET switches to the first node.
 18. The RF front-end ofclaim 15, wherein the capacitive element comprises one or morecapacitors.
 19. The RF front-end of claim 14, wherein the switch sizeweighting scheme is same as the capacitance weighting scheme.
 20. The RFfront-end of claim 15, wherein the second node is coupled to ground.